Adaptive radio transceiver with floating MOSFET capacitors

ABSTRACT

An exemplary embodiment of the present invention described and shown in the specification and drawings is a transceiver with a receiver, a transmitter, a local oscillator (LO) generator, a controller, and a self-testing unit. All of these components can be packaged for integration into a single IC including components such as filters and inductors. The controller for adaptive programming and calibration of the receiver, transmitter and LO generator. The self-testing unit generates is used to determine the gain, frequency characteristics, selectivity, noise floor, and distortion behavior of the receiver, transmitter and LO generator. It is emphasized that this abstract is provided to comply with the rules requiring an abstract which will allow a searcher or other reader to quickly ascertain the subject matter of the technical disclosure. It is submitted with the understanding that it will not be used to interpret or limit the scope or the meaning of the claims.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.10/796,817, filed Mar. 9, 2004, now U.S. Pat. No. 6,920,311, which is acontinuation of U.S. application Ser. No. 09/691,630 filed Oct. 18,2000, now U.S. Pat. No. 6,738,601, which is a continuation of U.S.application Ser. No. 09/634,552 filed Aug. 8, 2000.

U.S. application Ser. No. 09/634,552 claims priority to and claimsbenefit from U.S. Application Ser. No. 60/160,806 filed Oct. 21, 1999;U.S. Application Ser. No. 60/163,487 filed Nov. 4, 1999; U.S.Application Ser. No. 60/163,398 filed Nov. 4, 1999; U.S. ApplicationSer. No. 60/164,442 filed Nov. 9, 1999; U.S. Application Ser. No.60/164,194 filed Nov. 9, 1999; U.S. Application Ser. No. 60/164,314filed Nov. 9, 1999; U.S. Application Ser. No. 60/165,234 filed Nov. 11,1999; U.S. Application Ser. No. 60/165,239 filed Nov. 11, 1999; U.S.Aplication Ser. No. 60/165,356 filed Nov. 12, 1999; U.S. ApplicationSer. No. 60/165,355 filed Nov. 12, 1999; U.S. Application Ser. No.60/172,348 filed Dec. 16, 1999; U.S. Application Ser. No. 60/201,335filed May 2, 2000; U.S. Application Ser. No. 60/201,157 filed May 2,2000; U.S. Application Ser. No. 60/201,179 filed May 2, 2000; U.S.Application Ser. No. 60/202,997 filed May 10, 2000; and U.S. ApplicationSer. No. 60/201,330 filed May 2, 2000.

INCORPORATION BY REFERENCE

This application makes reference to the above-identified applicationswhich are hereby incorporated herein by reference in their entirety.

FIELD OF THE INVENTION

The present invention relates to telecommunication systems, and inparticular, to radio transceiver systems and techniques.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

[Not Applicable]

BACKGROUND OF THE INVENTION

Transceivers are used in wireless communications to transmit and receiveelectromagnetic waves in free space. In general, a transceiver comprisesthree main components: a transmitter, a receiver, and an LO generator orfrequency synthesizer. The function of the transmitter is to modulate,upconvert, and amplify signals for transmission into free space. Thefunction of the receiver is to detect signals in the presence of noiseand interference, and provide amplification, downconversion anddemodulation of the detected the signal such that it can be displayed orused in a data processor. The LO generator provides a reference signalto both the transmitter for upconversion and the receiver fordownconversion.

Transceivers have a wide variety of applications ranging from low datarate wireless applications (such as mouse and keyboard) to medium datarate Bluetooth and high data rate wireless LAN 802.11 standards.However, due to the high cost, size and power consumption of currentlyavailable transceivers, numerous applications are not being fullycommercialized. A simplified architecture would make a transceiver moreeconomically viable for wider applications and integration with othersystems. The integration of the transceiver into a single integratedcircuit (IC) would be an attractive approach. However, heretofore, theintegration of the transceiver into a single IC has been difficult dueto process variations and mismatches. Accordingly, there is a need foran innovative transceiver architecture that could be implemented on asingle IC, or alternatively, with a minimum number of discrete off-chipcomponents that compensate for process variations and mismatches.

SUMMARY OF THE INVENTION

In one aspect of the present invention, a capacitor having two nodesincludes a first transistor coupled to one of the two nodes, and asecond transistor coupled to the first transistor and to a second one ofthe two nodes.

In another aspect of the present invention, an integrated circuitincludes a capacitor having two nodes, a first transistor coupled to oneof the two nodes, and a second transistor coupled to the firsttransistor and to a second one of the two nodes.

In yet another aspect of the present invention, a tunable capacitorarray includes a plurality of capacitors each having first and secondnodes, a first transistor coupled to the first node, and a secondtransistor coupled to the second node, the first nodes of the capacitorsbeing coupled together and the second nodes of the capacitors beingcoupled together, and a plurality of switches each being positionedbetween a different one of the capacitors and the respective capacitorsfirst or second node.

It is understood that other embodiments of the present invention willbecome readily apparent to those skilled in the art from the followingdetailed description, wherein it is shown and described only embodimentsof the invention by way of illustration of the best modes contemplatedfor carrying out the invention. As will be realized, the invention iscapable of other and different embodiments and its several details arecapable of modification in various other respects, all without departingfrom the spirit and scope of the present invention. Accordingly, thedrawings and detailed description are to be regarded as illustrative innature and not as restrictive.

DESCRIPTION OF THE DRAWINGS

These and other features, aspects, and advantages of the presentinvention will become better understood with regard to the followingdescription, appended claims, and accompanying drawings where:

FIG. 1 is a block diagram of a transceiver in accordance with anexemplary embodiment of the present invention;

FIG. 2 is a block diagram of the transceiver blocks including areceiver, transmitter and local oscillator in accordance with anexemplary embodiment of the present invention;

FIG. 3 is a block diagram of a mixer in accordance with an exemplaryembodiment of the present invention;

FIG. 4 is an electrical diagram of a low noise amplifier in accordancewith an exemplary embodiment of the present invention;

FIG. 4( a) is an electrical diagram of a low noise amplifier inaccordance with an another exemplary embodiment of the presentinvention;

FIG. 5 is a block diagram of a four-stage biquad complex bandpass filterin accordance with an exemplary embodiment of the present invention;

FIG. 6 is an electrical diagram of one biquad stage of the complexbandpass filter in accordance with an exemplary embodiment of thepresent invention;

FIG. 7 is a graphical depiction of the frequency response on the biquadstage of FIG. 6 in accordance with an exemplary embodiment of thepresent invention;

FIG. 8 is an electrical diagram of one possible input circuit for thebiquad stage in accordance with an exemplary embodiment of the presentinvention;

FIG. 9 is an electrical diagram of another possible input circuit forthe biquad stage in accordance with an exemplary embodiment of thepresent invention;

FIG. 10 is an electrical diagram of a modified a modified biquad stageof FIG. 6 in accordance with an exemplary embodiment of the presentinvention;

FIG. 11 is a graphical depiction of the frequency response of themodified biquad stage of FIG. 10 in accordance with an exemplaryembodiment of the present invention;

FIG. 12( a) is an electrical diagram of a tunable array of capacitors inaccordance with an exemplary embodiment of the present invention;

FIG. 12( b) is an electrical diagram of ta tunable array of resistors inaccordance with an exemplary embodiment of the present invention;

FIG. 13 is a block diagram of a complex bandpass filter using polyphasein accordance with an exemplary embodiment of the present invention;

FIG. 14 is a block diagram of a programmable multiple stage amplifier inaccordance with an exemplary embodiment of the present invention;

FIG. 15 is a block diagram of an input and/or output stage for theprogrammable multiple stage amplifier of FIG. 14 in accordance with anexemplary embodiment of the present invention;

FIG. 16( a) is a block diagram of one core amplifier stage of theprogrammable multiple stage amplifier of FIG. 14 in accordance with anexemplary embodiment of the present invention;

FIG. 16( b) is a block diagram of a full-wave rectifier of the of theprogrammable multiple stage amplifier of FIG. 14 in accordance with anexemplary embodiment of the present invention;

FIG. 17( a) is an IF mixer in accordance with an exemplary embodiment ofthe present invention;

FIG. 17( b) is a graphical depiction of a frequency spectrum for thelimited IF clocks into the mixer of FIG. 17( a) in accordance with anexemplary embodiment of the present invention;

FIG. 17( c) is a graphical depiction of a frequency spectrum for the IFinput into the mixer of FIG. 17( a) in accordance with an exemplaryembodiment of the present invention;

FIG. 17( d) is a graphical depiction of a frequency spectrum for theoutput of the mixer of FIG. 17( a) in accordance with an exemplaryembodiment of the present invention;

FIG. 18 is a clock generator in accordance with an exemplary embodimentof the present invention;

FIG. 19( a) is a graphical depiction of a clock signal spectrum inputinto the clock generator of FIG. 18 in accordance with an exemplaryembodiment of the present invention;

FIG. 19( b) is a graphical depiction of a signal spectrum at the outputof a two second stage polyphase filter of the clock generator of FIG. 18in accordance with an exemplary embodiment of the present invention;

FIG. 19( c) is a graphical depiction of the signal spectrum output froma low pass filter of the clock generator of FIG. 18 in accordance withan exemplary embodiment of the present invention;

FIG. 20( a) is a graphical depiction of a signal spectrum at the inputto a polyphase filter in accordance with an exemplary embodiment of thepresent invention;

FIG. 20( b) is a graphical depiction of a signal spectrum at the outputof the polyphase filter in accordance with an exemplary embodiment ofthe present invention;

FIG. 20( c) is a graphical depiction of the signal spectrum output froma low pass filter of the polyphase filter in accordance with anexemplary embodiment of the present invention;

FIG. 21 is a block diagram of a demodulator in accordance with anexemplary embodiment of the present invention;

FIG. 22 is a block diagram of a differentiator of the demodulator ofFIG. 21 in accordance with an exemplary embodiment of the presentinvention;

FIG. 23 is a block diagram of a multiplier of the demodulator of FIG. 21in accordance with an exemplary embodiment of the present invention;

FIG. 24 is a block diagram of a peak detector/slicer of the demodulatorof FIG. 21 in accordance with an exemplary embodiment of the presentinvention;

FIG. 25 is a block diagram of a differential power amplifier inaccordance with an exemplary embodiment of the present invention;

FIG. 26( a) is a electrical diagram of one bias circuit to the inputand/or output stage of the differential power amplifier of FIG. 25 inaccordance with an exemplary embodiment of the present invention;

FIG. 26( b) is an electrical diagram of another bias circuit to theinput and/or output stage of the differential power amplifier of FIG. 25in accordance with an exemplary embodiment of the present invention;

FIG. 27 is an electrical diagram of a bias circuit for a the currentsource of the differential power amplifier of FIG. 25 in accordance withan exemplary embodiment of the present invention;

FIG. 28 is an electrical diagram of a power control circuit for thedifferential power amplifier of FIG. 25 in accordance with an exemplaryembodiment of the present invention;

FIG. 29 is an electrical diagram of a single-ended differential poweramplifier in accordance with an exemplary embodiment of the presentinvention;

FIG. 30 is an electrical diagram of digitally programmable CMOS poweramplifier in accordance with an exemplary embodiment of the presentinvention;

FIG. 31( a) is a block diagram of a local oscillator (LO) architecturein accordance with an exemplary embodiment of the present invention;

FIG. 31( b) is a block diagram of an LO architecture in accordance withanother exemplary embodiment of the present invention;

FIG. 32 is a block diagram of a LO architecture in accordance with analternative exemplary embodiment of the present invention;

FIG. 33 is a block diagram of a LO architecture in accordance with anyet another exemplary embodiment of the present invention;

FIG. 33( a) is a block diagram of a limiting buffer for the LOarchitecture of FIG. 33 in accordance with an exemplary embodiment ofthe present invention;

FIG. 34 is a block diagram of a wide tuning range voltage controlledoscillator (VCO) in accordance with an exemplary embodiment of thepresent invention;

FIG. 35 is an electrical diagram of the wide tuning range VCO of FIG. 34in accordance with an exemplary embodiment of the present invention;

FIG. 36( a) is a graphical depiction showing a typical VCO tuning curve;

FIG. 36( b) is a graphical depiction of a segmented VCO tuning curve inaccordance with an exemplary embodiment of the present invention;

FIG. 37( a) is a block diagram of a cross-coupled VCO in combinationwith a frequency divider in accordance with an exemplary embodiment ofthe present invention;

FIG. 37( b) is a block diagram of a VCO in combination with a dividerand polyphase circuit in accordance with an exemplary embodiment of thepresent invention;

FIG. 38 is a block diagram of a controller in accordance with anexemplary embodiment of the present invention;

FIG. 39 is an electrical diagram of an RC calibration circuit inaccordance with an exemplary embodiment of the present invention;

FIG. 40 is a block diagram of an RC calibration circuit using polyphasein accordance with an exemplary embodiment of the present invention;

FIG. 41 is an electrical diagram of a capacitor array in accordance withan exemplary embodiment of the present invention;

FIG. 42 is an electrical diagram of a bandgap calibration circuit inaccordance with an exemplary embodiment of the present invention;

FIG. 43 is a block diagram of bandgap circuit in accordance with anexemplary embodiment of the present invention;

FIG. 44 is a electrical diagram of a resistor array in accordance withan exemplary embodiment of the present invention;

FIG. 45 is a block diagram of a floating MOS capacitor in accordancewith an exemplary embodiment of the present invention;

FIG. 46 is an electrical diagram of a duplexing circuit with the poweramplifier on and the low noise amplifier off in accordance with anexemplary embodiment of the present invention; and

FIG. 47 is an electrical diagram of a duplexing circuit with the lownoise amplifier on and the power amplifier off in accordance with anexemplary embodiment of the present invention.

DETAILED DESCRIPTION

Exemplary Embodiments of a Transceiver

In accordance with an exemplary embodiment of the present invention, atranceiver utilizes a combination of frequency planning, circuit design,layout and implementation, differential signal paths, dynamiccalibration, and self-tuning to achieve robust performance over processvariation and interference. This approach allows for the fullintegration of the transceiver onto a single IC for a low cost, lowpower, reliable and more compact solution. This can be achieved by (1)moving external bulky and expensive image reject filters, channel selectfilters, and baluns onto the RF chip; (2) reducing the number ofoff-chip passive elements such as capacitors, inductors, and resistorsby moving them onto the chip; and (3) integrating all the remainingcomponents onto the chip. As those skilled in the art will appreciate,the described exemplary embodiments of the transceiver do not requireintegration into a single IC and may be implemented in a variety of waysincluding discrete hardware components.

As shown in FIG. 1, a described exemplary embodiment of the transceiverincludes an antenna 8, a switch 9, a receiver 10, a transmitter 12, alocal oscillator (LO) generator (also called a synthesizer) 14, acontroller 16, and a self-testing unit 18. All of these components canbe packaged for integration into a single IC including components suchas filters and inductors.

The transceiver can operate in either a transmit or receive mode. In thetransmit mode, the transmitter 12 is coupled to the antenna 8 throughthe switch 9. The switch 9 provides sufficient isolation to preventtransmitter leakage from desensitizing or damaging the receiver 10. Inthe receive mode, the switch 9 directs signal transmissions from theantenna 8 to the receiver 10. The position of the switch 9 can becontrolled by an external device (not shown) such as a computer or anyother processing device known in the art.

The receiver 10 provides detection of desired signals in the presence ofnoise and interference. It should be able extract the desired signalsand amplify it to a level where information contained in the receivedtransmission can be processed. In the described exemplary embodiment,the receiver 10 is based on a heterodyne complex (I-Q) architecture witha programmable intermediate frequency (IF). The LO generator 14 providesa reference signal to the receiver 10 to downconvert the receivedtransmission to the programmed IF.

A low IF heterodyne architecture is chosen over a direct conversionreceiver because of the DC offset problem in direct conversionarchitectures. DC offset in direct conversion architectures arises froma number of sources including impedance mismatches, variations inthreshold voltages due to process variations, and leakage from the LOgenerator to the receiver. With a low IF architecture, AC couplingbetween the IF stages can be used to remove the DC offset.

The transmitter 12 modulates incoming data onto a carrier frequency. Themodulated carrier is upconverted by the reference signal from the LOgenerator 14 and amplified to a sufficient power level for radiationinto free space through the antenna 8. The transmitter uses a directconversion architecture. With this approach only one step ofupconversion is required This leads to a reduction in both circuitcomplexity and power consumption.

The controller 16 performs two functions. The first function providesfor adaptive programming of the receiver 10, transmitter 14 and LOgenerator 16. By way of example, the transceiver can be programmed tohandle various communication standards for local area networks (LAN) andpersonal area networks (PAN) including HomeRF, IEEE 802.11, Bluetooth,or any other wireless standard known in the art. This entailsprogramming the transceiver to handle different modulation schemes anddata rates. The described exemplary embodiment of the transceiver cansupport modulation schemes such as Binary Phase Shift Keying (BPSK),Quadrature Phase Shift Keying (QPSK), offset quadrature phase shiftkeying (OQPSK), Multiple frequency modulations such as M level frequencyshift keying (FSK), Continuous Phase Frequency Shift Keying modulation(CFSK), Minimum Shift Keying modulation (MSK), Gaussian filtered FSKmodulation (GFSK), and Gaussian filtered Minimum Shift Keying (GMSK),Phase/Amplitude modulation (such as Quadrature Amplitude Modulation(QAM)), orthogonal frequency modulation (such as Orthogonal FrequencyDivision Multiplexing (OFDM)), direct sequence spread spectrum systems,and frequency hopped spread spectrum systems and numerous othermodulation schemes known in the art. Dynamic programming of thetransceiver can also be used to provide optimal operation in thepresence of noise and interference. By way of example, the IF can beprogrammed to avoid interference from an external source.

The second function provides for adaptive calibration of the receiver10, transmitter 14 and LO generator 16. The calibration functionalitycontrols the parameters of the transceiver to account for process andtemperature variations that impact performance. By way of example,resistors can be calibrated within exacting tolerances despite processvariations in the chip fabrication process. These exacting tolerancescan be maintained in the presence of temperature changes by adaptivelyfine tuning the calibration of the resistors.

The controller 16 can be controlled externally by a central processingunit (CPU), a microprocessor, a digital signal processor (DSP), acomputer, or any other processing device known in the art. In thedescribed exemplary embodiment, a control bus 17 provides two waycommunication between the controller 16 and the external processingdevice (not shown). This communication link can be used to externallyprogram the transceiver parameters for different modulation schemes,data rates and IF operating frequencies. The output of the controller 16is used to adjust the parameters of the transceiver to achieve optimalperformance in the presence of process and temperature variations forthe selected modulation scheme, data rate and IF.

The self-testing unit 18 generates test signals with differentamplitudes and frequency ranges. The test signals are coupled to thereceiver 10, transmitter 12 and LO generator 14 where they are processedand returned to the self-testing unit 18. The return signals are used todetermine the gain, frequency characteristics, selectivity, noise floor,and distortion behavior of the receiver 10, transmitter 12 and LOgenerator 14. This is accomplished by measuring the strength of thesignals output from the self-testing unit 18 against the returnedsignals over the tested frequency ranges. In an exemplary embodiment ofthe self-testing unit 18, these measurements can be made with differenttransceiver parameters by sweeping the output of the controller 16through its entire calibrating digital range, or alternatively makingmeasurements with the controller output set to a selected few points, byway of example, at the opposite ends of the digital range.

In the described exemplary embodiment, the self-testing unit 18 is incommunication with the external processing device (not shown) via thecontrol bus 17. During self-test, the external processing deviceprovides programming data to both the controller 16 and the self-testingunit 18. The self-testing unit 18 utilizes the programming data used bythe controller 16 to set the parameters of the transceiver to determinethe gain, frequency characteristics, selectivity, noise floor, anddistortion behavior of the receiver 10, transmitter 12 and LO generator14.

FIG. 2 shows a block diagram of the transceiver in accordance with anembodiment of the invention. The described exemplary embodiment isintegrated into a single IC. For ease of understanding, each componentcoupled to the controller is shown with a “program” designation or a“calibration” designation. These designations indicate whether thecomponent is programmed by the controller or calibrated by thecontroller. In practice, in accordance with the described exemplaryembodiment of the present invention, the components that are programmedreceive the MSBs and the components that are calibrated receive theLSBs. The components requiring both programming and calibration receivethe entire digital output from the controller. As those skilled in theart will appreciate, any number of methodologies may be used to deliverprogramming and calibration information to the individual components. Byway of example, a single controller bus could be used having theprogramming and or calibration data with the appropriate componentaddresses.

The receiver 10 front end includes a low noise amplifier (LNA) 22 whichprovides high gain with good noise figure performance. Preferably, thegain of the LNA 22 can be set by the controller (not shown) through a“select gain” input to maximize the receivers dynamic range. Thedesirability of dynamic gain control arises from the effect of blockersor interferers which can desensitize the LNA. Conventional filterdesigns at the input of the LNA 22 may serve to sufficiently attenuateundesired signals below a certain power level, however, for higher powerblockers or interferers, the LNA 22 should be operated with low gain.

The output of the LNA 22 is downconverted to a low IF frequency by thecombination of complex IF mixers 24 and a complex bandpass filter 26.More particularly, the output of the LNA 22 is coupled to the complex IFmixers 24 which generate a spectrum of frequencies based upon the sumand difference of the received signal and the RF clocks from the LOgenerator. The complex bandpass filter passes the complex IF signalwhile rejecting the image of the received signal. The image rejectioncapability of the complex IF mixers 24 in cooperation with the complexbandpass filter 26 eliminates the need for the costly and powerconsuming preselect filter typically required at the input of the LNAfor conventional low IF architectures.

The output of the complex bandpass filter 26 is coupled to aprogrammable multiple gain stage amplifier 28. The amplifier 28 can bedesigned to be programmable to select between a limiter and an automaticgain control (AGC) feature, depending on the modulation scheme used inthe transceiver. The limiting amplifier can be selected if thetransceiver uses a constant envelope modulation such as FSK. AGC can beselected if the modulation is not a constant envelope, such as QAM. Inaddition, the bandwidth of the amplifier 28 can be changed by thecontroller to accommodate various data rates and modulation schemes.

The output of the amplifier 28 is coupled to a second set of complex IFmixers 30 where it is mixed with the IF clocks from the LO generator forthe purpose of downconverting the complex IF signal to baseband. Thecomplex IF mixers 30 not only reject the image of the complex IF signal,but also reduces some of the unwanted cross modulation spurious signalsthereby relaxing the filtering requirements.

The complex baseband signal from the mixers 30 is coupled to aprogrammable passive polyphase filter within a programmable low passfilter 32. The programmable low pass filter 32 further filters outhigher order cross modulation products. The polyphase filter can becentered at four times the IF frequency to notch out one of the majorcross modulation products which results from the multiplication of thethird harmonic of the IF signal with the IF clock. After the complexbaseband signal is filtered, it either is passed through ananalog-to-digital (A/D) converter 34 to be digitized or is passed to ananalog demodulator 36. The analog demodulator 36 can be implemented tohandle any number of different modulation schemes by way of example FSKEmbodiments of the present invention with an FSK demodulator uses theA/D converter 36 to sample baseband data with other modulation schemesfor digital demodulation in a digital signal processor (not shown).

The LO generator 14 provides the infrastructure for frequency planning.The LO generator 14 includes an IF clock generator 44 and an RF clockgenerator 47. The IF clock generator includes an oscillator 38 operatingat a ratio of the RF signal (f_(OCS)) High stability and accuracy can beachieved in a number of ways including the use of a crystal oscillator.

The reference frequency output from the oscillator 38 is coupled to adivider 40. The divider 40 divides the reference signal f_(OSC) by anumber L to generate the IF clocks for downconverting the complex IFsignal in the receiver to baseband. A clock generator 41 is positionedat the output of the divider 40 to generate a quadrature sinusoidalsignal from the square wave output of the divider 40. Alternatively, theclock generator 41 can be located in the receiver. The divider 40 may beprogrammed by through the program input. This feature allows changes inthe IF frequency to avoid interference from an external source.

The output of the divider 40 is coupled to the RF clock generator 47where it is further divided by a number n by a second divider 42. Theoutput of the second divider 42 provides a reference frequency to aphase lock loop (PLL) 43. The PLL includes a phase detector 45, a divideby M circuit 46 and a voltage controlled oscillator (VCO) 48. The outputof the VCO 48 is fed back through the divide by M circuit 46 to thephase detector 45 where it is compared with the reference frequency. Thephase detector 45 generates an error signal representative of the phasedifference between the reference frequency and the output of the divideby M circuit 46. The error signal is fed back to the control input ofthe VCO 48 to adjust its output frequency f_(VCO) until the VCO 48 locksto a frequency which is a multiple of the reference frequency. The VCO48 may be programmed by setting M via the controller through the programinput to the divide by M circuit 46. The programmability resolution ofthe VCO frequency f_(VCO) is set by the reference frequency which alsomay be programmed by the controller through the program input of thedivider 42.

In the described exemplary embodiment, the VCO frequency is sufficientlyseparated (in frequency) from the RF frequency generated by thetransmitter 12 to prevent VCO pulling and injection lock of the VCO.Transmitter leakage can pull the VCO frequency toward the RF frequencyand actually cause the VCO to lock to the RF signal if their frequenciesare close to each other. The problem is exasperated if the gain andtuning range of the VCO is large. If the frequency of the RF clocks isf_(LO), then the VCO frequency can be defined as: f_(VCO)=Nf_(LO)/(N+1).This methodology is implemented with a divide by N circuit 50 coupled tothe output of the VCO 48 in the PLL 43. The output of the VCO 48 and theoutput of the divide byN circuit 50 are coupled to a complex mixer 52where they are multiplied together to generate the RF clocks. A filter53 can be positioned at the output of the complex mixer to remove theharmonics and any residual mixing images of the RF clocks. The divide byN circuit can be programmable via the controller through the selectinput. For example, if N=2, then f_(VCO)=(⅔)f_(LO), and if N=3, thenf_(VCO)=(¾) f_(LO).

A VCO frequency set at ⅔ the frequency of the RF clocks works well inthe described exemplary embodiment because the transmitter output issufficiently separated (in frequency) from the VCO frequency. Inaddition, the frequency of the RF clocks is high enough so that itsharmonics and any residual mixing images such as f_(VCO)×1−(1/N)), 3f_(VCO)×1+(1/N), and 3 f_(VCO)×1−(1/N)) are sufficiently separated (infrequency) from the transmitter output to relax the filteringrequirements of the RF clocks. The filtering requirements do not have tobe sharp because the filter can better distinguish between the harmonicsand the residual images when they are separated in frequency.Programming the divide by N circuit 50 also provides for the quadratureoutputs of the divide by N circuit. Otherwise, with an odd numberprogrammed, the outputs of the divide by N circuit 50 would not bequadrature. For an odd number, the divider 50 outputs will bedifferential, but will not be 90 degrees out of phase, i.e., will not beI-Q signals.

In the described exemplary embodiment, the RF clocks are generated inthe in the LO generator 14. This can be accomplished in various fashionsincluding, by way of example, either generating the RF clocks in the VCOor using a polyphase circuit to generate the RF clocks. Regardless ofthe manner in which the RF clocks are generated, the mixer 52 willproduce a spectrum of frequencies including the sum and differencefrequencies, specifically, f_(VCO)×(1+(1/N)) and its imagef_(VCO)×(1−(1/N)). To reject the image, the mixer 52 can be configuredas a double quadrature mixer as depicted in FIG. 3. The doublequadrature mixer includes one pair of mixers 55, 57 to generate theQ-clock and a second pair of mixers 59, 61 to generate the I-clock. TheQ-clock mixers utilizes a first mixer 55 to mix the I output of the VCO48 (see FIG. 2) with the Q output of the divider 40 and a second mixer57 to mix the Q output of the VCO with the I output of the divider. Theoutputs of the first and second mixers are connected together togenerate the Q-clock. Similarly, the I-clock mixers utilizes a firstmixer 59 to mix the I output of the divider with the Q output of the VCOand a second mixer 61 to mix the Q output of the divider with the Ioutput of the VCO. The outputs of the first and second mixers areconnected together to generate the I-clock. This technique provides veryaccurate I-Q clocks by combination of quadrature VCO and filtering.Because of the quadrature mixing, the accuracy of the I-Q clocks is notaffected by the VCO inaccuracy, provided that the divide by N circuitgenerates quadrature outputs. This happens for even divide ratios, suchas N=2.

Optimized performance is achieved through frequency planning andimplemented by programmable dividers in the LO generator to selectdifferent ratios. Based on FIG. 2, all the dependencies of thefrequencies are shown by the following equation:f _(LO) =f _(RF)−(M×f _(OSC) /nL)(1+1/N)=f _(OSC) /L

where f_(RF) is frequency of the transmitter output.

Turning back to FIG. 2, the transmitter 12 includes a complex buffer 54for coupling incoming I-Q modulated baseband signals to a programmablelow-pass filter 56. The low-pass filter 56 can be programmed by thecontroller through the select input. The output of the low-pass filter56 is coupled to complex mixers 58. The complex mixers 58 mixes the I-Qmodulated baseband signals with the RF clocks from the LO generator todirectly upconvert the baseband signals to the transmitting frequency.The upconverted signal is then coupled to an amplifier 60 and eventuallya power amplifier (PA) 62 for transmission into free space through theantenna A bandpass filter (not shown) may be disposed after the PA 62 tofilter out unwanted frequencies before transmission through the antenna.

In the described exemplary embodiment, the transmitter can be configuredto minimize spurious transmissions. Spurious transmissions in a directconversion transmitter are generated mainly because of the nonlinearityof the complex mixers and the DC offsets at the input to the complexmixers. Accordingly, the complex mixers can be designed to meet aspecified IIP3 (Input Intercept Point for the 3^(rd) Harmonic) for themaximum allowable spurs over the frequency spectrum of thecommunications standard. The DC offsets at the input to the complexmixers can be controlled by the physical size of the transistors.

In addition, the transmitter can be designed to minimize spurioustransmission outside the frequency spectrum of the communicationsstandard set by the FCC. There are two sources for these spurs: the LOgenerator and the transmitter. These spurs can be are suppressed bymultiple filtering stages in the LO generator and transmitter.Specifically, in the LO generator, due to the complex mixing of the VCOsignal with the output of the divide by N circuit, all the spurs are atleast f_(VCO)/N away from the RF clocks. By setting N to 2, by way ofexample, these unwanted spurs will be sufficiently separated (infrequency) from the transmitted signal and are easily removed byconventional filters in the LO generator and transmitter. Thus, thespurs will be mainly limited to the harmonics of the transmitted signal,which are also sufficiently separated (in frequency) from thetransmitted signal, and therefore, can be rejected with conventionalfiltering techniques. For further reduction in spurs, a dielectricfilter may be placed after the PA in the transmitter.

1.0 Receiver

1.1 Differential Amplifier

In exemplary embodiments of the present invention, a differentialamplifier can be used to provide good noise immunity in low noiseapplications. Although the differential amplifiers are described in thecontext of a low noise amplifier (LNA) for a transceiver, those skilledin the art will appreciate that the techniques described are likewisesuitable for various applications requiring good noise immunity.Accordingly, the described exemplary embodiments of an LNA for atransceiver is by way of example only and not by way of limitation.

1.1.1 Single-to-Differential LNA

The described LNA can be integrated into a single chip transceiver orused in other low noise applications. In the case of transceiver chipintegration, the LNA should be relatively insensitive to the substratenoise or coupling noise from other transceiver circuits. This can beachieved with a single-to-differential LNA. The single-ended inputprovides an interface with an off-chip single-ended antenna. Thedifferential output provides good noise immunity due to its common moderejection.

FIG. 4 shows a schematic of a single-to-differential amplifier havingtwo identical cascode stages that are driven by the same single-endedinput 64. The input 64 is coupled to a T-network having two seriescapacitors 82, 84 and a shunt inductor 72. The first stage includes apair of transistors 74, 78 connected between the shunt inductor 72 and aDC power source via an inductor 68. The second stage includes acomplimentary pair of transistors 76, 80 connected between ground andthe DC power source via an inductor 70. The gate of the one of thetransistors 80 in the second stage is connected to the output of theT-network at the capacitor 84. A bias current is applied to the gate ofeach transistor.

This configuration provides an input that is well matched with theantenna because the parallel connection of the T-network with the sourceof the transistor 78 transforms the 1/gm (transconductance) of thetransistor to a resistance (preferably 50 ohms to match the antenna). Byadjusting the values of T-network components, the matching circuit canbe tuned for different frequencies and source impedances. The inputcapacitor 82 of the T-network further provides decoupling between theantenna and the amplifier.

For DC biasing purposes, the shunt inductor 72 provides a short circuitto ground allowing both stages of the amplifier to operate at the sameDC drain current. The output capacitor 84 provides DC isolation betweenthe gate bias applied to the transistor 80 of the second stage and thesource 82 of the transistor 78 in the first stage.

In operation, a signal applied to the input of the amplifier is coupledto both the source 82 of the transistor 78 of the first stage and thegate 83 of the transistor 80 of the second stage. This causes the gainof each stage to vary inversely to one another. As a result, the signalvoltage applied to the input of the amplifier is converted to a signalcurrent with the signal current in the first stage being inverted fromthe signal current in the second stage. Moreover, the two stages willgenerate the same gain because the gm of the transistors should be thesame, and therefore, the total gain of the amplifier is twice as much asconventional single-to-differential amplifiers.

1.1.2 Differential LNA

A differential LNA can also be used to provide good noise immunity inlow noise applications, such as the described exemplary embodiment ofthe transceiver. In FIG. 4( a), an exemplary differential LNA is shownhaving a cascode differential pair with inductive degeneration. In thedescribed exemplary embodiment, the differential LNA can be integratedinto a single chip transceiver or used in other similar applications.

In the case of transceiver chip integration, an off chip coupler (notshown) can be used to split the single-ended output from the antennainto a differential output with each output being 180° out of phase. TheLNA input can be matched to the coupler, i.e., a 50 ohm source, by LCcircuits. A shunt capacitor 463 in combination with a series inductor465 provides a matching circuit for one output of the coupler, and ashunt capacitor 467 in combination with a series inductor 469 provides amatching circuit for the other output of the coupler. At 2.4 GHz., eachLC circuit may be replaced by a shunt capacitor and transmission line.In the described exemplary embodiment, the LC circuits are off-chip forimproved noise figure performance. Alternatively, the LC circuits couldbe integrated on chip. However, due to the high loss of on chipinductors, the noise figure, as well as gain, could suffer.

The differential output of the coupler is connected to a differentialinput of the LNA via the LC matching circuits. The differential inputincludes a pair of input FET transistors 471,473 with inductivedegeneration. This is achieved with an on chip source inductor 475connected between the input transistor 471 and ground, and a second onchip source inductor 479 connected between the input transistor 473 andground. The on chip inductive degeneration provides a predominantlyresistive input impedance. In addition, the FET noise contribution atthe operating frequency is reduced.

The outputs of the input transistors 471, 473 are coupled to a cascodestage implemented with a pair of transistors 481, 486, respectively. Thecascode stage provides isolation between the LNA input and its output.This methodology improves stability, and reduces the effect of theoutput load on the LNA input matching circuits. The gates of the cascodetransistors 481, 486 are biased at the supply voltage by a resistor 488.The resistor 488 reduces instability that might otherwise be caused byparasitic inductances at the gates of the cascoded transistors 481, 486.Since the described exemplary embodiment of the LNA uses a differentialarchitecture, the resistor does not contribute noise to the LNA output.

The output of cascoded transistor 481 is coupled to the supply voltagethrough a first inductor 490. The output of the cascodedc transistor 486is coupled to the supply voltage through a second inductor 492. The LNAis tuned to the operating frequency by the output inductors 490, 492.More particularly, these inductors 490, 492 resonate with the LNA outputparasitic capacitance, and the input capacitance of the next state (notshown). Embodiments of the present invention integrated into a singleintegrated circuit do not require a matching network at the LNA output.

The gain of the LNA can be digitally controlled. This is achieved byintroducing a switchable resistor in parallel with each of the outputinductors. In the described exemplary embodiment, a series resistor 494and switch 496 is connected in parallel with the output inductor 490,and a second series resistor 498 and switch 500 is connected in parallelwith the output inductor 492. The switches can be FET transistors or anyother similar switching devices known in the art. In the low gain mode,each resistor 494, 498 is connected in parallel with its respectiveoutput inductor 490, 492, which in turn, reduces the quality factor ofeach output inductor, and as a consequence the LNA gain. In the highgain mode, the resistors 494, 498 are switched out of the LNA outputcircuit by their respective switches 496, 500.

1.2 A Complex Filter

In an exemplary embodiment of the present invention, aprogrammable/tunable complex filter is used to provide frequencyplanning, agility, and noise immunity. This is achieved with variablecomponents to adjust the frequency characteristics of the complexfilter. Although the complex filter is described in the context of atransceiver, those skilled in the art will appreciate that thetechniques described are likewise suitable for various applicationsrequiring frequency agility or good noise immunity. Accordingly, thedescribed exemplary embodiment for a complex filter in a transceiver isby way of example only and not by way of limitation.

The described complex filter can be integrated into a single chiptransceiver or used in other low noise applications. In the case oftransceiver chip integration, the off-chip filters used for imagerejection and channel selection can be eliminated. A low-IF receiverarchitecture enables the channel-select feature to be integrated intothe on-chip filter. However, if the IF lies within the bandwidth of thereceived signal, e.g. less than 80 MHz in the Bluetooth standard, theon-chip filter should be a complex filter (which in combination with thecomplex mixers) can suppress the image signal. Thus, either a passive oran active complex filter with channel select capability should be used.Although a passive complex filter does not dissipate any power byitself, it is lossy, and loads the previous stage significantly. Thus,an active complex filter with channel select capability is preferred.The channel select feature of the active complex filter can achievecomparable performance to conventional band-pass channel-select filtersin terms of noise figure, linearity, and power consumption.

The described exemplary embodiment of the complex filter accommodatesseveral functions in the receiver signal path: it selects the desiredchannel, rejects the image signal which lies inside the data band of thereceived signal due to its asymmetric frequency response, and serves asa programmable gain amplifier (PGA). Moreover, the complex filter centerfrequency and its bandwidth can be programed and tuned. Thesecapabilities facilitate a robust receiver in a wireless environment,where large interferers may saturate the receiver or degrade thesignal-to-noise ratio at the demodulator input. The attenuation of thereceived signal at certain frequencies can also be enhanced byintroducing zeros in the complex filter.

1.2.1 Cascaded Biquads

An exemplary embodiment of the complex filter includes a cascade ofbiquads. Each biquad comprises a 2'nd order bandpass filter. The totalorder of the filter is the sum of orders of the cascaded biquads. Theorder of the filter can be programmable. By way of example, fourcascaded biquads 83, 85, 87, 89 can be used with each of the cascadedbiquads having an individually controlled bypass switch. Referring toFIG. 5, a bypass switch 91 is connected across the input stage biquad83. Similarly, a bypass switch 93 is connected across the second stagebiquad 85, a bypass switch 95 is connected across the third stage biquad87, and a bypass switch 97 is connected across the output stage biquad.With this configuration, the order of the filter can be programmed bybypassing one or more biquads. A biquad that is bypassed contributes azero order to the filter.

In the described exemplary embodiment, the bypass switches are operatedin accordance with the output from the controller 16 (see FIG. 2). An8'th order filter can be constructed by opening the bypass switches 91,93, 95, 97 via the digital signal from the controller output. Thecomplex filter can be reduced to a 6'th order filter by closing thebypass switch 97 to effectively remove the output stage biquad from thecomplex filter. Similarly, the complex filter can be reduced to a 4'thorder filter by closing bypass switches 95, 97 effectively removing thethird stage biquad and output stage biquad. A 2'nd order filter can becreated by closing bypass switches 93, 95, 97 effectively removing allbiquads with the exception of the input stage from the circuit.

1.2.1.1 The Poles of a Biquad Stage

FIG. 6 shows an exemplary embodiment of a biquad stage of the complexfilter. The biquad stage includes two first order resistor-capacitor(RC) filters each being configured with a differential operationalamplifier 94, 96, respectively. The first differential operationalamplifier 94 includes two negative feedback loops, one between eachdifferential output and its respective differential input. Each feedbackloop includes a parallel RC circuit (98, 106), (108, 100), respectively.Similarly, the second differential operational amplifier 96 includes twonegative feedback loops, one between each differential output and itsrespective differential input. Each feedback loop includes a parallel RCcircuit (102-110), (112-104), respectively. This topology is highlylinear, and therefore, should not degrade the overall IIP3 of thereceiver. The RC values determine the pole of the biquad stage.

The differential inputs of the biquad stage are coupled to theirrespective differential operational amplifiers through input resistors114, 116, 118, 120. The input resistors in combination with theirrespective feedback resistors set the gain of the biquad stage.

Preferably, some or all of the resistors and capacitors values canprogrammable and can be changed dynamically by the controller Thismethodology provides a frequency agile biquad stage.

The two first order RC filters are cross coupled by resistors 86, 88,90, 92. By cross-coupling between the two filters, a complex responsecan be achieved, that is, the frequency response at the negative andpositive frequencies will be different. This is in contrast to areal-domain filter, which requires the response to be symmetric at bothpositive and the negative frequencies. This feature is useful becausethe negative frequency response corresponds to the image signal. Thus,the biquad stage selects the desired channel, whereas the image signal,which lies at the negative frequency is attenuated.

For the resistor values shown in FIG. 6, the biquad stage outputs are:

$\begin{matrix}{V_{OI} = {A\frac{{\left( {1 + {j\;{RC}\;\omega}} \right)V_{II}} + {2{QV}_{IQ}}}{\left( {1 + {j\;{RC}\;\omega}} \right)^{2} + {4Q^{2}}}\mspace{14mu}{and}}} & (1) \\{V_{OQ} = {A\frac{{{- 2}{QV}_{II}} + {\left( {1 + {j\;{RC}\;\omega}} \right)V_{IQ}}}{\left( {1 + {j\;{RC}\;\omega}} \right)^{2} + {4Q^{2}}}}} & (2)\end{matrix}$

FIG. 7 shows the frequency response for the complex biquad filter.

After the received signal is downconverted, the desired channel in the Ipath lags the one in the Q path, that is, V_(1I)=−jV_(1Q), andtherefore:

$\begin{matrix}{{H({j\omega})} = {{\frac{V_{o}}{V_{I}}({j\omega})} = \frac{A}{1 + {j\;{RC}\;\omega} - {j2Q}}}} & (3)\end{matrix}$

This shows a passband gain of A 122 at a center frequency of 2 Q/RC 124,with a 3-dB bandwidth of 2 RC 126. Thus, the quality factor of thesecond-order stage will be Q. For the image signal however, the signalat the I branch leads, and as a result:

$\begin{matrix}{{H({j\omega})} = \frac{A}{1 + {j\;{RC}\;\omega} + {j2Q}}} & (4)\end{matrix}$which shows that the image located at 2 Q/RC is rejected by

$\frac{1}{\sqrt{\left( {1 + {4Q}} \right)2}}.$Therefore, the biquad stage has an asymmetric frequency response, thatis, the desired signal may be assigned to positive frequencies, whereasthe image is attributed to negative frequencies. In general, thefrequency response of the biquad stage is obtained by applying thefollowing complex-domain transformation to a normalized real-domainlowpass filter:

$\begin{matrix}\left. {j\omega}\rightarrow\frac{j\left( {\omega - \omega_{0}} \right)}{BW} \right. & (5)\end{matrix}$where ω₀ is the bandpass (BP) center frequency, and BW is the lowpass(LP) equivalent bandwidth, equal to half of the bandpass filterbandwidth. For instance, for a second-order biquad stage (as shown inFIG. 6), ω₀=2 Q/RC, and BW=1/RC. The biquad stage is designed by findingits LP equivalent frequency response using equation (5). Once the LPpoles are known, the BP poles are calculated based on equation (5).Assume that the LP equivalent has n poles, and p_(i,LP)=αi+jβ_(i) is theith pole. From equation (5), the BP pole will be:P _(i,BP) =BW·P _(i,LP) +jω ₀=α_(i) ·BW+j(ω₀+β_(i) BW)  (6)

The complex filter is realized by cascading n biquad stages. Therefore,similar to real-domain bandpass filters, an nth order complex filteruses 2×n integrators. Based on equation (3), each biquad stage has apole equal to −1/RC+j2 Q/RC. Thus:

$\begin{matrix}{{\alpha_{i} \cdot {BW}} = {\frac{- 1}{RC}\mspace{14mu}{and}}} & (7) \\{{\omega_{o} + {\beta_{i} \cdot {BW}}} = \frac{2Q}{RC}} & (8)\end{matrix}$

Since the LP equivalent poles are located in the left-half plane, a_(i)is always negative. The above equations set the value of Q and RC ineach stage. The gain of each biquad stage can be adjusted based on thedesired gain in the complex filter, and noise-linearity trade-off:increasing the gain of one biquad stage lowers the noise contributed bythe following biquad stages, but it also degrades the linearity of thecomplex filter.

In addition to image rejection, the complex frequency transformation ofthe biquad stage (equation (5)) provides for its frequency response tobe symmetric around its center frequency as shown in FIG. 7. This is incontrast to regular bandpass filters which use the following real-domaintransformation:

$\begin{matrix}\left. {j\omega}\rightarrow\frac{j\left( {\omega^{2} - \omega_{0}^{2}} \right)}{{BW} \cdot \omega} \right. & (9)\end{matrix}$

This symmetric response in the biquad stage ensures a uniform groupdelay across the data band.

1.2.1.2 The Zeros of a Biquad Stage

The described exemplary embodiment of the biquad stage can be modifiedto obtain a sharper rejection or notch at an undesired signal at aspecific frequency. This can be achieved in the biquad stage by addingzeros. Assume that the input resistors at the biquad input (R_(i) 114 inFIG. 6) is replaced with an admittance Y_(i). For the received signal,the frequency response of the biquad stage will be equal to:

$\begin{matrix}{{H({j\omega})} = \frac{R \cdot Y_{i}}{1 + {j\;{RC}\;\omega} - {j2Q}}} & (10)\end{matrix}$

FIG. 8 shows Yi having resistor R_(z) 128 and capacitor C_(z) 130.

In order to have a zero located at jω axis in the frequency response,Y_(i) should contain a term such as 1−ω/ω_(z). If Y_(i) is simply madeof a resistor R_(z) in parallel with a capacitor C_(z), then the inputadmittance will be equal to:

$\begin{matrix}{Y_{i} = {\frac{1}{R_{z}} + {{j\omega}\; C_{z}}}} & (11)\end{matrix}$which is not desirable, since the zero will be in the left-half plane,rather than the j ω axis.

FIG. 9 shows Yi with the capacitor C_(z) 132 connected to the Q input134 and the resistor R_(Z) connected to the I input 136. Now the currentI will be equal to:

$\begin{matrix}{I = {\frac{V}{R_{z}} + {j\; C_{z}{\omega \cdot ({jV})}}}} & (12)\end{matrix}$

Therefore, the input admittance will be equal to:

$\begin{matrix}{Y_{i} = {\frac{1}{V} = {\frac{1}{R_{z}} - {C_{z}\omega}}}} & (13)\end{matrix}$which indicates that the filter will have a zero equal to 1/R_(z) C_(z)at the jω axis.

FIG. 10 shows a single biquad stage modified to have a zero at the jωaxis. The biquad stage includes capacitors 138, 140, 142, 144. Thecombination of capacitors 138, 140, 142, 148 and resistors 116, 118determines a complex zero with respect to the center frequency. Thetransfer function for the received signal will be:

$\begin{matrix}{{H\left( {j\;\omega} \right)} = {A\frac{1 - {\frac{R\; C_{z}}{A}\omega}}{1 + {j\; R\; C\;\omega} - {{j2}\; Q}}}} & (14)\end{matrix}$

Equation (14) is analogous to equation (3), with the difference that nowa zero at A/RC_(z) is added to the biquad stage of the complex filter.By knowing the LP equivalent characteristics of the biquad stage, thepoles are calculated based on equation (6). The value of Q and RC ineach biquad stage is designed by using equation (7) and equation (8). Ifthe normalized LP zeros are at ±ω_(z,LP), then the biquad stage shouldbe realized with two biquad stages cascoded, and the frequency of zerosin the biquad stages will be (equation (5)):ω_(z1,2)=ω₀±ω_(z,LP) ·BW  (15)

If the differential I and Q inputs connected to the zero capacitors areswitched, the biquad stage will have zeros at negative frequencies(image response). This property may be exploited to notch the imagesignal.

1.2.1.3 Tunability and Programmability

In addition to channel selection and image rejection, the describedexemplary embodiment of the complex filter can provide variable gain,bandwidth, and center frequency. In addition, an automatic tuning loopcan be implemented to adjust the center frequency. These features resultin a high quality receiver which can dynamically support differentcommunication standards, modulation schemes and data rates.

By changing the gain of the biquad stages, the complex filter canperform as a PGA in the signal path of the receiver. This assures thatthe output swing of the complex filter remains constant when thereceiver input signal changes. Moreover, adaptivity is achieved throughdynamic programming of the bandwidth and center frequency. By way ofexample, when the receive environment is less noisy, the transmitter mayswitch to a higher data rate, and the bandwidth of the complex filtershould increase proportionally. The center frequency, on the other hand,may be changed to increase the receiver immunity to blockers and otherinterferers.

The center frequency of each biquad stage is equal to 2 Q/RC. Thequality factor, Q, is precisely set, since it is determined by the ratioof two resistors (R_(f) and R_(c) in FIG. 10), which can be accuratelyestablished when the resistors are implemented on-chip. However, the RCproduct varies by temperature and process variations, and therefore, maybe compensated by automatic tuning methods.

Referring to FIG. 12( a), each capacitor can be implemented with acapacitor 148 connected in parallel with a number of switchablecapacitors 150, 152, 154, 156. The capacitance, and thereby the centerfrequency of the complex filter, can be varied by selectively switchingin or out the capacitors based on a four-bit binary code. Each bit isused to switch one of the parallel capacitors from the circuit In thedescribed exemplary embodiment, the capacitor 148 provides a capacitanceof C_(u)/2. Capacitor 150 provides a capacitance of C_(u)/2. Capacitor152 provides a capacitance of C_(u)/4. Capacitor 154 provides acapacitance of C_(u)/8. Capacitor 156 provides a capacitance ofC_(u)/16. This provides ±50% tuning range with ±3% tuning accuracy. Dueto discrete nature of the tuning scheme, there may be some error in thecenter frequency (±1/(2×2^(n)) for n-bit array). This inaccuracy can betolerated with proper design.

Referring to FIG. 12( b), each resistor can be implemented with a seriesof switchable resistors 158, 160, 162, 164, 166. Resistor 166 provides aresistance of R_(u). Resistor 164 provides a resistance of 2 R_(u).Resistor 162 provides a resistance of 4 R_(u). Resistor 160 provides aresistance of 8 R_(u). Resistor 158 provides a resistance of 16 R_(u).In the described exemplary embodiment, the resistance can be variedbetween R_(u) and 31×R_(u) in incremental steps equal to R_(u) byselectively bypassing the resistor based on a five-bit binary code.

The center frequency of the complex filter can be adjusted by setting1/R_(u)C_(u) equal to a reference frequency generated, by way ofexample, the crystal oscillator in the controller. The filter isautomatically tuned by monotonic successive approximation as describedin detail in Section 4.0 herein. Once the value of R_(u) C_(u) is set,the complex filter characteristics depends only on four-bit code for thecapacitors and the four-bit code for the resistors. For example, assumethat the value of the resistors in the biquad stage of FIG. 6 is asfollowing: R_(i)=n_(A) R_(u), R_(f)=n_(Q) R_(u), and Rc=n_(Q) R_(u).Likewise, assume that C=n_(C) C_(u), where n_(C) is a constant, and that1/R_(u) C_(u)=ω_(u). The value of ω_(u) is set to a reference crystal bya successive approximation feedback loop. The filter frequency responsefor the received signal will be:

$\begin{matrix}{{H\left( {j\;\omega} \right)} = \frac{\frac{n_{F}}{n_{A}}}{1 + {j\; n_{c}n_{F}R_{u}C_{u}\omega} - {j\frac{n_{F}}{n_{Q}}}}} & (16)\end{matrix}$

Therefore, the biquad stage gain (A), center frequency (ω₀), andbandwidth (BW) will be equal to:

$\begin{matrix}{A = \frac{n_{F}}{n_{A}}} & (17) \\{\omega_{0} = {\frac{1}{n_{C}n_{Q}} \cdot \omega_{u}}} & (18) \\{{B\; W} = {\frac{1}{n_{C}n_{F}} \cdot \omega_{u}}} & (19)\end{matrix}$

The above equations show that the characteristics of the biquad stage isindependently programmed by varying n_(A), n_(F), and n_(Q). Forinstance, by setting n_(F), the gain of the biquad stage changes fromn_(F)/31 to n_(F) by changing n_(A) from 1 to 31.

1.2.2 I-Q Monolithic Bandpass Filter

Alternatively, a low power I-Q monolithic bandpass filter can be usedfor the complex filter of the described exemplary embodiment of thepresent invention. The I-Q monolithic bandpass filter is useful forshort-range communication applications. It also provides low powermonolithic bandpass filtering for high data rates such as Bluetooth andHomeRF applications. The I-Q monolithic bandpass filter can be fullyincorporated in monolithic channel select filters for 1-MHz data rates.

FIG. 13 is a block diagram of the I-Q monolithic bandpass filter inaccordance with an embodiment of the present invention. The I-Qmonolithic bandpass filter includes a cascode of selectively intertwinedbiquads 168 and polyphase circuits 170. The biquads can be the same asthe biquads described in Section 1.2.1 herein, or any other biquadsknown in the art. Similarly, the polyphase circuits can also be anyconventional polyphase circuits known in the art. The biquad circuitscan be 2'nd order lowpass filters, which in conjunction with thepolyphase circuits, exhibit a 1-MHz bandwidth bandpass filter with morethan 45 dB rejection for all frequencies beyond 2 MHz away from thecenter of the band. The number of biquads determines the order of theI-Q monolithic bandpass filter. The polyphase filters are for widerbandwidth and image rejection. The number of polyphase filtersdetermines the number of zeros in the frequency response of the I-Qmonolithic bandpass filter.

In the described embodiment, an 8'th order Butterworth filter isimplemented in conjunction with selective side band filtering ofpolyphase circuits to create a low IF I-Q monolithic bandpass filter.The described embodiment of the I-Q monolithic bandpass filter does notsuffer excessive group delay despite large bandwidth. The input IP3 canbe better than 5 dBm with a gain of more than 20 dB and the noise figurecan be less than 40 dB. In fully integrated embodiments of the presentinvention, the I-Q monolithic bandpass filter can have on chip tuningcapability to adjust for process, temperature and frequency variations.

1.3 Programmable Multiple Gain Amplifier

In one exemplary embodiment of the present invention, a programmablemultiple gain amplifier is used in the receiver path between the complexfilter and the complex IF mixer (see FIG. 2). The programmable multiplegain amplifier can be designed to be programmable to select between alimiter and an AGC feature. The programmable multiple gain amplifier,when operating as a limiter provides a maximum gain for frequencymodulation applications. The programmable multiple gain amplifier,operating as an AGC, can be used for applications utilizing amplitudemodulation.

FIG. 14 shows a block diagram of an exemplary embodiment of theprogrammable multiple gain amplifier with an RSSI output. The RSSIoutput provides an indication of the strength of the IF signal. Theprogrammable multiple gain amplifier includes three types of amplifiers.The input buffer is shown as a type I amplifier 900 and the type IIIamplifier 904 serves as the output buffer. The core amplifier is shownas a direct-coupled cascade of seven differential amplifiers 930, 931,932, 933, 934, 935, 936. The core amplifier includes seven bypassswitches 930′, 931′, 932′, 933′, 934′, 935′, 936′, one bypass switchconnected across each differential amplifier. The bypass switchesprovide programmable gain under control of the controller (see FIG. 2).

When the programmable gain amplifier is operating as a limiter, all thebypass switches will be opened by the controller. Conversely, when theprogrammable gain amplifier is operating in the AGC mode, the outputgain of the core amplifier will be varied by controlling the bypassswitch positions to prevent saturation of the core amplifier by largesignals. In the described exemplary embodiment, the RSSI signal is fedback to control the bypass switch positions through a digital AGC loopin the external processing device. The AGC loop provides information tothe controller 16 via the control bus 17 regarding the optimum gainreduction (see FIG. 2). The controller translates the information fromthe external processing device into a digital signal for controlling thebypass switch positions of the core amplifier accordingly. The largerthe RSSI signal, the greater the gain reduction of the core amplifierwill be and the more bypass switches that will be closed by thecontroller.

In one embodiment of the programmable gain amplifier, the type I andtype III amplifiers can be the same. FIG. 15 shows one possibleconstruction of these amplifiers. In this configuration, transistors952, 954 provide amplification of the differential input signal. Thedifferential input signal is fed to the gates of transistor amplifiers952, 954, and the amplified differential output signal is taken from thedrains. The gain of the transistor amplifiers 952, 954 is set by loadresistors 956, 958. Transistors 960, 962 provide a constant currentsource for the transistor amplifiers 952, 954. The load resistors 956,958, connected between the drain of their respective transistoramplifiers 952, 954 and a common gate connection of transistors 960,962, provides a bias current source to common mode feedback.

Turning back to FIG. 14, the type II core amplifier 902 includes adirect-coupled cascade of seven differential amplifiers 930, 931, 932,933, 934, 935, 936, each with a voltage gain, by way of example, 12 dB.The voltage at the output of each differential amplifier 930, 931, 932,933, 934, 935, 936 is coupled to a rectifier 937, 938, 939, 940, 941,942, 943, 944, respectively. The outputs of the rectifiers are connectedto ground through a common resistor 945. The summation of the currentsfrom each of the rectifiers flowing through the common resistor providesa successive logarithmic approximation of the input IF voltage. With a12 dB gain per each differential amplifier, a total cascaded gain of 84dB is obtained. As those skilled in the art will appreciate, any numberof differential amplifiers, each with the same or different gain, may beemployed.

The input dynamic range of an RSSI is explained using the followingderivation. Throughout this section, assume each rectifier has an idealsquare law characteristic and its transfer function is:y=β ² V _(in) ²  (20)

Now, assume that S is the maximum input range of one differentialamplifier and rectifier combination, whichever is smaller. This isdetermined with the lowest of the two values V_(i) and V_(L) that arethe maximum input range of each differential amplifier, and the maximuminput range of the rectifier, respectively.S=min(V _(i) ,V _(L))  (21)

Therefore, the RSSI maximum input level is S, and the ideal RSSI minimuminput level is S/A^(n), where A is the gain of each differentialamplifier and n is the number of the differential amplifiers. Thus, theideal dynamic range is calculated as follows:

$\begin{matrix}{{{Ideal}\mspace{14mu}{Dynamic}\mspace{14mu}{Range}} = {{20\;\log\frac{S}{\frac{S}{A^{n}}}} = {{20\;\log\; A^{n}} = {20(n)\;\log\; A}}}} & (22)\end{matrix}$

However, in the case of a large amount of gain, the input level will belimited with the input noise and the dynamic range will also be limitedto:

$\begin{matrix}\begin{matrix}{{{Dynamic}\mspace{14mu}{Range}} = {20\;\log\frac{S}{\sqrt{\sigma_{n}}}}} \\{\sqrt{\sigma_{n}} = {{total}\mspace{14mu}{noise}\mspace{14mu}{rms}}} \\{\sigma_{n} = {({BW}) \times {Noise}\mspace{14mu}{Factor}}}\end{matrix} & (23)\end{matrix}$

If each differential amplifier has the same input dynamic range V_(L)and each full-wave rectifier has similar input dynamic range V_(i), thenthe dynamic range of the logarithmic differential amplifier and thetotal RSSI circuitry are the same.

The logarithmic approximations are provided by piecewise linearsummation of the rectified output of each differential amplifier. Thisis done by segmentation of the input voltage by the power of 1/A.Successively, each differential amplifier will reach the limiting pointas the input signal grows by the power of A. Assuming each rectifier ismodeled as shown in equation (20), the logarithmic approximation ismodeled as following:

For an input being in the following range:

$\begin{matrix}{\frac{S}{A^{n - m}} < V_{i\; n} < \frac{S}{A^{n - m - 1}}} & (24)\end{matrix}$up to the last m stages of the differential amplifier are all beinglimited and the rest of the differential amplifiers are in the lineargain region. Therefore, the RSSI is shown to be:A ²β²ν_(in) ² +A ⁴β⁴ν_(in) ⁴ + . . . +A ^(2(n−m))β^(2(n−m))ν_(in)^(2(n−m)) +mβ² S ² =RSSI  (25)

This is further simplified to:

$\begin{matrix}{{RSSI} = {{\frac{\left( {A\;\beta} \right)^{2}}{\left( {A\;\beta} \right)^{2} - 1}{V_{i\; n}^{2}\left\lbrack {\left( {A\;\beta} \right)^{2{({n - m - 1})}} -} \right\rbrack}} + {m\;\beta^{2}S^{2}}}} & (26) \\{{RSSI} \approx {{\frac{1}{\left( {A\;\beta} \right)^{2} - 1}{V_{i\; n}^{2}\left( {A\;\beta} \right)}^{2{({n - m})}}} + {m\;\beta^{2}S^{2}}}} & (27)\end{matrix}$

The above equation is a first order approximation to the logarithmicfunction shown in equation (28) according to the first two terms of theTaylor expansion at a given operating point.Ideal RSSI=C log V _(in) ²  (28)

The following calculates the constant C from the maximum and minimum ofthe RSSI:Max RSSI−Min RSSI=C logA^(2n)  (29)ΔRSSI=C logA^(2n)  (30)

$\begin{matrix}{C = \frac{\Delta\;{RSSI}}{2\; n\;\log\; A}} & (31) \\{{({Ideal})\mspace{11mu}{RSSI}} = {\frac{\Delta\mspace{11mu}{RSSI}}{2\; n\;\log\; A}\log\; V_{i\; n}^{2}}} & (32)\end{matrix}$

To find the relation between the gain of a differential amplifier, thegain of a rectifier, and the maximum input range of the combineddifferential amplifier and the rectifier, the RSSI will be calculatedfor the two consecutive differential amplifier and rectifiercombinations (see equations (33) and (34)) for both ideal RSSI equations(32) and approximated RSSI equation (27):

$\begin{matrix}{V_{i\; n\; 1} = \frac{S}{(A)^{n - m}}} & (33) \\{V_{i\; n\; 2} = \frac{S}{(A)^{n - m - 1}}} & (34)\end{matrix}$(Ideal)RSSI ₂ −RSSI ₁=log(A)²  (35)(Approximated)RSSI ₂ −RSSI ₁=β²S²  (36)

Therefore,Clog(A)²=β² S ⁶²  (37)

Using equations (18) and (12), the following expression is achieved:

$\begin{matrix}{\frac{\Delta\;{RSSI}}{n} = {\beta^{2}S^{2}}} & (38)\end{matrix}$

Plugging equation (19) into (8) results in the following:

$\begin{matrix}{{{RSSI} = {{\frac{1}{\left( {A\;\beta} \right)^{2} - 1}\left( {A\;\beta} \right)^{2{({n - m})}}V_{in}^{2}} + {m\;\frac{\Delta\;{RSSI}}{n}}}};{\frac{S}{A^{n - m}} < V_{in} < \frac{S}{A^{n - m - 1}}}} & (39)\end{matrix}$

FIG. 16( a) shows a schematic diagram for an exemplary embodiment of thedifferential amplifier used in the type II core amplifier. Thedifferential input signal is fed to the gates of transistor amplifiers955, 957. The amplified differential output signal is provided at thedrains of the transistor amplifiers 955, 957. The gain of the transistoramplifiers is set by load transistors 958, 860, each connected betweenthe drain of one of the transistor amplifiers and a power source. Moreparticularly, the gain of the differential amplifier is determined bythe ratio of the square root of transistor amplifiers-to-loadtransistors.

$\begin{matrix}\begin{matrix}{{{Gain}(A)} = \sqrt{\frac{w_{in}}{w_{in}}}} \\{= \sqrt{\frac{200}{6} \approx 5.8}}\end{matrix} & (40)\end{matrix}$The sources of the transistor amplifiers 955, 957 are connected incommon and coupled to a constant current source transistor 952. In thedescribed exemplary embodiment, the controller provides the bias to thegate of the transistor 952 to set the current.

An exemplary embodiment of the full-wave rectifier with two unbalancedsource-coupled pairs cross-coupled is shown in FIG. 16( b). In thisembodiment, the differential input signal is fed to an unbalanced pairof transistors. One of the differential input pairs is fed to the gatesof the unbalanced transistor pair 968, 966 and the other differentialinput pair is fed to the gates of the other unbalanced transistor pair964,962. The drains of transistors 968,962 are connected in common andprovide one of the differential output pairs. The drains of transistors964, 966 are connected in common and provide the other differentialoutput pair. Transistors 968,964 are connected in a common sourceconfiguration and coupled to a constant current source transistor 965.Transistors 962, 966 are also connected in a common source configurationwith the common source connected to a current source transistor 967. Thegates of the current sources 965, 967 are connected together. In thedescribed exemplary embodiment, the controller provides the bias to thecommon gate connection to set the current.

Transistors 970 and 971 provide a current-mirror load to cross-coupledtransistors 968, 962. Similarly, transistors 972, 973 provide acurrent-mirror load to cross-coupled transistors 962, 964. The currentthrough the cross-coupled transistors 962, 964 is the sum of the currentthrough the load transistor 972 and the current through the loadtransistor 971 which is mirrored from the load transistor 970. Thecurrent through the cross-coupled transistors 962, 962 is also mirroredto load transistor 973 for the RSSI output.

When the transistors 962, 964, 966, and 968 are operating in thesaturation region, the following equations are shown for thedifferential output current DI_(SQB1) where k is the ratio of the twounbalanced source-coupled transistors:

$\begin{matrix}\begin{matrix}{{{if}\mspace{14mu}\Delta\; I_{SQM1}} = {\left( {I_{D\; 1} + I_{D\; 4}} \right) - \left( {I_{D\; 2} + I_{D\; 3}} \right)}} \\{= {2\left( {I_{DC} + I_{SQ}} \right)}} \\{= {{2\frac{k - 1}{k + 1}I_{o}} - {4\frac{k\left( {k - 1} \right)\beta_{N}}{\left( {k + 1} \right)^{2}}V_{I}^{2}}}}\end{matrix} & (41)\end{matrix}$

The input dynamic range of the full rectifier is then:

$\begin{matrix}{{{{if}\mspace{14mu}\Delta\; I_{SQM1}} = O},{V_{i} = {{\pm \frac{\sqrt{I_{o}}}{\beta_{N}}}\frac{\sqrt{k + 1}}{2k}}}} & (42)\end{matrix}$

The full-wave rectifier includes two unbalanced differential pairs witha unidirectional current output. One rectifier 976 taps eachdifferential pair and sums their currents into a 10 kW resistor R_(L).

The square law portion of equation (41) multiplied by the resistanceprovides the β²S² of equation (42):

$\begin{matrix}{{\beta^{2}S^{2}} = {4\frac{k\left( {k - 1} \right)\beta_{N}}{\left( {k + 1} \right)^{2}}V_{i}^{2}R_{L}}} & (43)\end{matrix}$

By plugging the V_(i) from equation (42) and replacing β²S² fromequation (38), the following relation is obtained:

$\begin{matrix}{\frac{\Delta\;{RSSI}}{n} = {2\frac{k - 1}{k + 1}I_{o}R_{L}}} & (44)\end{matrix}$

For ΔRSSI=1V, n=7 stages, R_(L)=10000Ω, and k=4, from the above equationI_(o) is calculated to be 12 mA. Therefore, each rectifier will bebiased with two 12 mA current sources (one 12 ma current source for theI signal and a second 12 ma current source for the Q channel). Thisresults in an approximately logarithmic voltage, which indicates thereceived signal-strength (RSSI).

1.4 Complex IF Mixers

The IF down conversion to baseband signal can be implemented using fourfully balanced quadrature mixers as shown in FIG. 17( a). This mixerconfiguration includes both quadrature inputs from the programmablemultiple stage amplifier and quadrature IF clocks from the LO generator.This configuration produces single sideband, quadrature basebandsignals, with minimum number of spurs at the output. Thesecharacteristics aid in relaxing the baseband filtering as well assimplifying the demodulator architecture. An IF mixer buffer 352 buffersthe IF clock (Clk_I, Clk_Q as shown in FIG. 17( a)).

The outputs of the limiters are coupled to the quadrature clocks of theIF mixers (I_in for mixer 322, I_in for mixer 323, Q_in for mixer 324,Q_in for mixer 325) and the IF clocks are coupled to the data input ofthe IF mixers. This configuration minimizes spurs at the output of theIF mixers because the signal being mixed is the IF clocks which is aclean sine wave, and therefore, has minimal harmonics. The limitingaction of the programmable multiple stage amplifier on the I and Q datawill have essentially no effect on the spurs at the output of the IFmixers. FIG. 17 b shows the IF mixer clock signal spectrum whichcontains only odd harmonics. The IF signals do not have even harmonicsin embodiments of the present invention using a fully differentialconfiguration. The bandwidth of the m'th(=2 n+1) harmonic is directlyproportional to mfs, whereas its amplitude is inversely proportional tomfs. FIG. 17 c shows the sinusoidal input spectrum of the IF clocks.FIG. 17 d shows the IF mixer output spectrum.

1.5 Clock Generator

A clock generator can be used to generate a quadrature sinusoidal signalwith controlled amplitude. The clock generator can be located in thereceiver, or alternatively the LO Generator, and provides a cleansinusoidal IF from the square wave output of the divider in the LOGenerator for downconverting the IF signal in the receiver path tobaseband. FIG. 18 shows a block diagram and signal spectrum of a clockgenerator. A sinusoidal signal is generated from a square-wave usingcascaded polyphase. FIG. 18 shows a clock generator block diagram. Theclock generator outputs clk_I and clk_Q for the IF mixer buffer (seeFIG. 17). The clock generator includes a polyphase filter at 3 fs 360, apolyphase filter at 5 fs 362, and a low pass filter 364. FIG. 19 a showsthe input clock signal spectrum. FIG. 19 b shows the spectrum at 3 fs366 and at 5 fs 368 polyphase. FIG. 19 c shows the sinusoidal signalgeneration after the low pass filter 364.

In fully integrated embodiments of the present invention, the controllercan provide self calibration to generate precise signal levels withnegligible dependency on the process variations. The two polyphasefilters 360,362 with RC calibration can be used to remove the first twoodd harmonics of the signal. The remaining harmonics can be filteredwith an on chip tunable low pass filter. The output of the clockgenerator block is a quadrature sinusoidal signal with controlled signallevel. This spectrally clean signal is used at the input of complex IFmixers to downconvert the IF signal to baseband.

1.6 Programmable Low Pass Filter

The first major spurs out of downconversion process is at 4 times the IFfrequency. A self calibrated 4 fs polyphase filter can be used after thecomplex IF mixers to reduce the spurious and improve the linearity ofthe demodulator.

The polyphase filter can be implemented with two back to back polyphaseto reject both positive phase and the negative phase. Built-inprogrammability can also be included for operating with otherfrequencies. This capability enables the demodulator to be highlyflexible. It can support wide range of incoming IF frequencies and withdifferent modulation schemes.

Following the polyphase filter, a quadrature lowpass filter can be usedto remove unwanted spurs. The lowpass filter can be programmable anddesigned to minimize group delay distortion without sacrificing highfrequency filtering characteristics.

In fully integrated embodiments of the present invention, the controllercan provide on chip RC calibration to minimize any process variation.The programmability of the polyphase filter and the low pass filter addsa new degree of flexibility to the system; it can be used to accommodatedifferent data bandwidths.

FIG. 20 shows a baseband spectrum filtering before the discriminator.FIG. 20( a) shows the signal spectrum at polyphase input, i.e., thefrequency spectrum of the polyphase filter. FIG. 20( b) shows the signalspectrum at polyphase output, i.e. the frequency spectrum of the lowpass filter. FIG. 20( c) shows the signal spectrum at the low passfilter output.

1.7 High Data Rate Frequency Demodulator

The demodulator may take on various forms to accommodate differentmodulation schemes. One embodiment of the demodulation used inconnection with the present invention includes a low power, monolithicdemodulator for high data rates in frequency modulated systems. Thisdemodulator can provide data recovery for well over 1-MHz data rates.

The demodulator can be FSK or GMSK demodulator. FSK is digital frequencymodulation. GMSK is a specific type of FSK. GMSK stands for Gaussianfiltered FSK modulation, which means that GMSK has gaussian filtering atthe output of frequency modulation. GMSK has more stringent requirementsthan FSK. The data rate is higher for GMSK and the modulation index islow for GMSK relative to FSK.

The described embodiment of the demodulator is a low power, fullyintegrated FSK/GMSK demodulator for high data rates and low modulationindex. The FSK operates with the programmable gain stage amplifier as alimiter, and therefore, does not require oversampling clocks or complexAGC blocks.

FIG. 21 is a block diagram of an exemplary high data rate frequencydemodulator in accordance with the present invention. The demodulatorperforms a balanced quadrature demodulation. Differentiators 329, 330convert the baseband signal to a signal having an amplitude proportionalto the baseband signal frequency. One differentiator 329 converts the Isignal and the other differentiator 330 converts the Q signal. The Isignal output of the differentiator 329 is coupled to a multiplier 331where it is multiplied by the Q signal input into the demodulator. The Qsignal output of the differentiator 330 is coupled to a multiplier 332where it is multiplied by the I signal input into the demodulator. Themultipliers 331, 332 produce a single ended DC signal. The DC signalsare summed together by summation circuit 333. A peak detector/slicer 334digitizes the DC signal from the summation circuit, thereby producingdiscrete zeros and ones.

The frequency discrimination can be performed using a differentiator asshown in FIG. 22. A differential input signal is coupled to the input ofan amplifier 340 through capacitors 341, 342. A feedback resistor 343,344 is coupled between each differential output. Its operation is basedon generating an output signal level linearly proportional to theincoming signal frequency. In other words, the higher the incomingfrequency, the larger signal amplitude output by the differentiator.Therefore, it is desirable to have a spur free signal at the input ofthis stage. High frequency spurs can degrade the performance of thedifferentiator. By using the polyphase filter in conjunction with thelowpass filter (see FIG. 2) before the demodulator, a nearly idealbaseband signal is input to the differentiator. The capacitors 341, 342in the signal path with the resistive feedback operation of theamplifier is proportional to the time derivative of the input. For asinusoidal input, V(in)=A.sin(ωt), the output will be V(out):d/dt(V(in))=to.A.cos(ωt). Thus, the magnitude of the output increaseslinearly with increasing frequency.

The controller provides RC calibration to keep the differentiation gainprocess invariant. In order to reduce the effect of any high frequencycoupling to the differentiator input, the differentiator gain isflattened out for frequencies beyond the band of interest. In additionto frequency discrimination, the differentiation process adds a 90degrees phase shift to the incoming signal. This phase shift is inherentto differentiation process. Since the output is in quadrature phase withthe input (except for differing amplitude), cross multiplication of theinput and output results in frequency information.

FIG. 23 shows an exemplary analog multiplier 331, 332 with zero higherharmonics in accordance with the present invention. Buffers one 334 andtwo 335 are added to a Gilbert cell to linearize the voltage levels.Buffers one 334 and two 335 convert the two inputs into two voltagelevels for true analog multiplication using a Gilbert cell. The Gilbertcell is comprised of transistors 336, 338, resistors 340, 342 andcross-coupled pairs of transistors 344, 346 and transistors 348, 350.

By cross multiplying the input and the output signals to thedifferentiator, the amplitude information is generated. Since thesignals are at baseband, it can be difficult to filter out any spursresulting from the multiplication process. Linearized buffers can beused to minimize spurs by providing a near ideal analog multiplier. Onchip calibration can also be used to control the multiplication gain andto minimize process variation dependency. In order to accommodate highdata rates such as 1 MHz and beyond, all the stages should have lowphase delays. In addition, matching all the delays in quadrature signalscan be advantageous.

The output of the multiplier is a single ended DC signal which is alinear function of the frequency. This analog output can representmultilevel FSK with arbitrary modulation index. The minimum modulationindex is only limited by wireless communication fundamentals.

An exemplary peak detector/slicer for frequency data detection is shownin FIG. 24. The differential input signal is coupled to a peak detector346 which detects the high peak. The differential input signal is alsocoupled to a second peak 347 detector which detects the low valley ofthe signal. The outputs of the peak detectors are coupled to a resistordivider network 348, 349 to obtain the average of the output signal. Theaverage signal output from the resistor divider network is used as thecalibrated zero frequency to obviate frequency offset problems due tothe frequency translation process from IF to baseband.

A differential amplifier 345 is used to digitize the frequencyinformation by comparing the differential input signal with thecalibrated zero frequency. The output of the amplifier is a logic “1” ifthe baseband frequency is greater than the calibrated zero frequency anda logic “0” if the baseband frequency is less than the calibrated zerofrequency. The output is amplified through several inverters 350 whichin turn generate digital rail to rail output.

2.0. Transmitter

2.1 Differential Power Amplifier

In an exemplary embodiment of the invention, the PA is a differential PAas shown in FIG. 25. The symmetry of the differential PA in conjunctionwith other features supports implementation in a variety of technologiesincluding CMOS. The described embodiment of the differential PA can be afully integrated class A PA. A balun 610 is used to connect the PA to anantenna or a duplexer. The balun converts the differential signal to asingle-ended output.

The described embodiments of the differential PA is a two stage device.The two stages minimize backward leakage of the output signal to theinput stage. As those skilled in the art will appreciate, any number ofstages can be implemented depending on the particular application andoperating environment. Equal distribution of gain between the two stageshelps prevents oscillation by avoiding excess accumulation of gain inone stage. A cascode architecture may be incorporated into the PA toprovide good stability and insulation.

The input stage or pre-amplifier of the power amplifier includes aninput differential pair comprising amplifying transistors 612, 614.Transistor 616 is a current source that biases the input differentialpair. The presence of a current source provides many positive aspectsincluding common mode rejection. The current is controlled by thevoltage applied to the gate of transistor 616. The gate voltage shouldbe chosen to prevent the transistor 616 from operating in the trioderegion. Triode operation of transistor 616 has a number of drawbacks.Primarily, since transistor 616 is supposed to act as a current source,its operation in the triode region can cause distortion in the currentflowing into the transistor 612 and the transistor 614, and consequentlygives rise to nonlinearity in the signal. Secondly, the triode behaviorof transistor 616 will depend on temperature and process variations.Therefore, the circuit operation will vary over different process andtemperature corners.

Cascode transistors 618, 620 provide stability by isolating the outputfrom the input. As a result, no change in the input impedance occursover frequency. The gates of the cascode transistors 618, 620 devicesare biased through a bond wire. A resistor 622 in series with the gatesof the cascode transistors prevents the inductance associated with thebonding from resonating with the input capacitive of the transistors,thereby improving stability. The resistor 622 in combination with thegates of transistors 618, 620 also improves common mode rejection andmakes the transistor input act like a virtual ground at RF. Resistor 623isolates the power supply from the PA and provides common mode rejectionby increasing the symmetry of the differential PA. Inductors 624, 626tune out the capacitance at the drains of the transistors 618, 620. Atthe tuning frequency, the impedance seen at the drains of thetransistors 618, 620 is high, which provides the high gain at the tuningfrequency.

The differential output of the input stage is provided at the drains ofthe cascode transistors 618, 620 to AC coupling capacitors 628, 630.Capacitor 628 couples the drain of transistor 618 with the gate oftransistor 632. Capacitor 630 couples the drain of transistor 620 withthe gate of transistor 634. The transistors 632, 634 provideamplification for the second stage of the PA. Resistors 636, 638 arebiasing resistors for biasing the transistors 632, 634.

In the output stage of the PA, the current level is higher and the sizeof the current source should be increased to maintain the same biassituation. However, large tail devices can lower the common moderejection. Accordingly, instead of a current source, an inductor 640 canbe used to improve the headroom. The inductor 640 is a good substitutefor a current source. The inductor 640 is almost a short circuit at lowfrequencies and provides up to 1 Kohm of impedance at RF. By way ofexample, a 15 nH inductor with proper shielding (to increase the Q) anda self-resonance frequency close to 4.5 GHz can be used for optimum highfrequency impedance and sufficient self-resonance.

Inductors 622, 624 tune out the capacitance at the drains of transistors632, 634. Capacitors 642, 644 are AC coupling capacitors. Inductors 646and capacitor 648 match the output impedance of the PA to the antenna,by way of example, 50 Ω. Similarly, inductors 650 and capacitor 652match the output impedance of the PA to the antenna. Balun 610 is adifferential to single-ended voltage converter. Resistance 654 isrepresentative of the load resistance.

Capacitances associated with bias resistors may also be addressed.Consider a typical distributed model for a polysilicon (“poly” forshort) resistor. Around 4 fF to substrate can be associated with everykilo-ohm of resistance in a poly resistor. This means that, for examplein a 20 Kohm resistor, around 80 fF of distributed capacitance to thesubstrate exists. This can contribute to power loss because part of thepower will be drained into the substrate. One way of biasing the inputstage and the output stage is through a resistive voltage divider asshown in FIG. 26( a). The biasing of the input stage is shown for thetransistor 616 in FIG. 25, however, those skilled in the art willreadily appreciate that the same biasing circuit can be used for thetransistor 614 (FIG. 25). One drawback from this approach, however, isthat the gate of the transistor will see the capacitance from the tworesistors 658, 660 of the voltage divider. Capacitor 662 is a couplingcapacitor, which couples the previous stage to the voltage divider.Switch 664 is for powering down the stage of the power amplifier that isconnected to the voltage divider. The switch 664 is on in normaloperation and is off in power down mode.

FIG. 26( b) is similar to FIG. 26( a), except that FIG. 26( b) includesresistor 666. DC-wise the FIG. 26( a) and FIG. 26( b) circuits are thesame. However, in AC, not only is the resistance seen from the gates oftransistors 634, 632 towards the resistive bias network bigger, but thecapacitance is smaller because the capacitance is caused by resistor 666and not resistors 660, 658. Since there is less capacitance, there isless loss of the signal. From FIG. 25, transistors 618, 620 in the inputstage and transistors 632, 634 in the output stage can be biased by theresistive voltage divider shown in FIG. 26( b).

FIG. 27 shows an exemplary bias circuit for the current sourcetransistor 616 of FIG. 25. To fix the bias current of the circuit overtemperature and process variation, a diode-connected switch transistor672 may be used with a well-regulated current 670. The voltage generatedacross the diode-connected transistor 672 is applied to the gate of thecurrent source transistor 616. Because of the mirroring effect of thisconnection and since all transistors move in the same direction overtemperature and process corners, the mirrored current will be almostconstant. The reference current is obtained by calibration of a resistorby the controller. The calibrated resistor can be isolated from the restof the PA to prevent high frequency coupling through the resistor toother transceiver circuits. As those skilled in the art will appreciate,the exemplary bias circuit is not limited to the current sourcetransistor of the PA and may be applied to other transistors requiringaccurate biasing currents.

FIG. 28 shows an exemplary power control circuit. The power controlcircuit can provide current scaling. The power control circuit changespower digitally by controlling the bias of the current source transistor616 of the first differential pair 612, 614 in the PA (FIG. 25). Thepower control circuit can be used in any application requiring differentpower levels. The power control is done by applying different voltagelevels to the gate of the current source in the first stage (input stageor preamplifier) of the PA. A combination of current adjustment in bothstages (input stage and output stage) of the PA can also be done.Different voltage levels are generated corresponding to different powerlevels. In one embodiment of the invention, the power control circuithas four stages as shown in FIG. 28. Alternatively, the power controlcircuit can have any number of stages corresponding to the number ofpower levels needed in an application.

The power control circuit includes transistor pairs in parallel.Transistors 674, 676, 678, 680 are switch transistors and are coupled todiode-connected transistors 682, 684, 686, 688, respectively. The switchtransistors 674, 676, 678, 680 are coupled to a current source 670. Eachdiode-connected transistor 682, 684, 686, 688 can be switched into theparallel combination of by turning its respective switching transistoron. Conversely, any diode-connected transistor can be removed from theparallel combination by turning its respective switch transistor off.The current from the current source 670 is injected into a parallelcombination of switch transistors 674, 676, 678, 680. The power levelcan be incremented or decremented by switching one or more switchtransistors into the parallel combination. By way of example, a decreasein the power level can be realized by switching a switch transistor intothe parallel combination. This is equivalent to less voltage drop acrossthe parallel combination, which in turn corresponds to a lower powerlevel. A variety of stages are comprehended in alternative embodimentsof the invention depending on the number of power levels needed for agiven application. A thermometer code from the controller can be appliedto the power control circuit according to which the power level isadjusted.

As described above, the output of the PA can be independently matched toa 50 ohm load. The matching circuit (inductors 646, 650 and capacitors648, 652) is connected to the balun. Any non-ideality of the balun, bondwire impedance, pin/PCB capacitance, and other parasitics can beabsorbed by the matching circuits. High-Q inductors can be used wherepossible. The loss in efficiency may also be tolerable with low powerapplications.

2.2 Single-Ended Differential Power Amplifier

In another embodiment of the present invention, the balun can beeliminated by a single-ended to differential PA. FIG. 29 shows theoutput stage of a single-ended to differential PA. The output stageincludes resistors 690, 692, inductors 694, 696, 698, and transistors700, 702. Coupling capacitor 704 couples the output stage to an LCcircuit, the LC circuit including inductor 706 and capacitor 708.Coupling capacitor 710 couples the second stage to a CL circuit, the CLcircuit comprising capacitor 712 and inductor 714. The transistors 700,702 provide amplification of the differential signal applied to theoutput stage of the PA. The output of the amplifying transistors 700,702 produces two signals 180 degrees out of phase. The LC circuit isused to match the first output to a 100 ohm load 718 and to shift thephase of the signal by 90 degrees. The CL circuit is deployed to matchthe second output to a 100 ohm load 720, and to shift the phase of thesignal in the opposite direction by 90 degrees. Since the two outputswere out of phase by 180 degrees at the beginning and each underwent anadditional 90 degrees of shift (in opposite directions) the two signalsappearing across the two 100 ohm loads will be in phase. In an idealsituation, they will also be of similar amplitudes. This means that thetwo nodes can be connected together to realize a single-ended signalmatched for a 50 ohm load 716.

Unlike the differential PA, the differential to single-endedconfiguration does not enjoy the symmetry of a fully differential path.Accordingly, with respect to embodiments of the present inventionintegrated into a single IC, the effect of bond wires should beconsidered. Because of stability and matching issues, a separate ground(bond wire) for the matching circuit should be used. The bond wiresshould be small and the matching should be tweaked to cancel theireffect.

The bias current to the amplifying transistors 700, 702 for embodimentsof the present invention integrated into a single IC can be set in anumber of ways, including by way of example, the bias circuit shown inFIG. 27. The voltage generated across the diode-connected transistor 672is applied to the gate of the amplifying transistor 700. A similar biascircuit can be used for biasing the amplifying transistor 702.

Alternatively, the bias circuit of the amplifying transistors 700, 702for single IC embodiments can be set with a power control circuit asshown in FIG. 28. The current source is connected directly theamplifying transistor 700. By incrementally switching thediode-connected transistors 682, 684, 686, 688 into the parallelcombination, the voltage applied to the gate of the amplifyingtransistor 700 is incrementally pulled down toward ground. Conversely,by incrementally switching the diode-connected transistors 682, 684,686, 688 out of the parallel combination, the voltage applied to thegate of the amplifying transistor 700 is incrementally pulled up towardthe source voltage (not shown). A similar power control circuit can beused with the amplifying transistor 702.

2.3. Digitally Programmable CMOS PA with On-Chip Matching

In another embodiment of the present invention, a PA is integrated intoa single IC with digitally programmable circuitry and on-chip matchingto an external antenna, antenna switch, or similar device. FIG. 30 showsan exemplary PA with digital power control. This circuit comprises twostages. The input stage provides initial amplification and acts as abuffer to isolate the output stage from the VCO. The output stage iscomprised of a switchable differential pair to steer the current towardsthe load. The output stage also provides the necessary drive for theantenna. The power level of the output stage can be set by individuallyturning on and off current sources connected to each differential pair.

Transistors 722, 724 provide initial amplification. Transistor 726 isthe current source that biases the transistors 722, 724. Inductors 728,730 tune out the capacitance at the drains of the transistors 722, 724.At the tuning frequency, the impedance seen at the drains is high, whichprovides high gain at the tuning frequency.

Capacitors 732, 736 are AC coupling capacitors. Capacitor 732 couplesthe drain of transistor 724 with the gate of transistor 734. Capacitor736 couples the drain of transistor 722 with the gate of transistor 738.Resistors 740, 742 are biasing resistors for biasing the gates of thetransistors 734, 738. Transistors 734, 738 are amplifying transistors inthe output stage of the PA. Transistor pairs 744, 746, transistor pairs748, 750, and transistor pairs 752, 754 each provide additional gain forthe signal. Each pair can be switched in or out depending on whether ahigh or low gain is needed. For maximum gain each transistor pair in theoutput stage of the PA will be switched on. The gain can beincrementally decreased by switching out individual transistor pairs.The PA may have more or less transistor pairs depending on the maximumgain and resolution of incremental changes in the gain that is desired.

Transistor 756 has two purposes. First, it is a current source thatbiases transistors 734, 738. Second, it provides a means for switchingtransistors 734, 738 in and out of the circuit to alter the gain of theoutput stage amplifier. Each transistors 758, 760, 762 serves the samepurpose for its respective transistor pair. A digital control, word fromthe controller can be applied to the gates of the transistors 756, 758,760, 762 to digitally set the power level. This approach provides theflexibility to apply ramp up and ramp down periods to the PA, inaddition to the possibility of digitally controlling the power level.The drains of the transistors 756, 758, 760, 762 are connected to acircuit that serves a twofold purpose: 1) it converts the differentialoutput to single ended output, and 2) it matches the stage to external50 ohm antenna to provide maximum transferable gain.

Inductors 764, 766 tune out the capacitance at the drains of transistors752, 754. Capacitor 768 couples the PA to the load 770. Inductor 772 isa matching and phase-shift element, which advances the phase of thesignal by 90°. Capacitor 794 is a matching and phase-shift element,which retards the phase of the signal by 90°. Capacitor 796 is the padcapacitance. The bonding wire 798 bonds the PA to the load resistance770 (e.g., the antenna).

3.0 Local Oscillator

In embodiments of the present invention utilizing a low-IF or directconversion architecture, techniques are implemented to deal with thepotential disturbance of the local oscillator by the PA. Since the LOgenerator has a frequency which coincides with the RF signal at thetransmitter output, the large modulated signal at the PA output may pullthe VCO frequency. The potential for this disturbance can be reduced bysetting the VCO frequency far from the PA output frequency. To this end,an exemplary embodiment of the LO generator produces RF clocks whosefrequency is close to the PA output frequency, as required in a low-IFor direct-conversion architectures, with a VCO operating at a frequencyfar from that of the RF clocks. One way of doing so is to use two VCO864, 866, with frequencies of f₁ and f₂ respectively, and mix 868 theiroutput to generate a clock at a higher frequency of f₁+f₂ as shown inFIG. 31( a). With this approach, the VCO frequency will be away from thePA output frequency with an offset equal to f₁ (or f₂). A bandpassfilter 876 after the mixer can be used to reject the undesired signal atf₁-f₂. The maximum offset can be achieved when f₁ is close to f₂.

An alternative embodiment for generating RF clocks far away in frequencyfrom the VCO is to generate f₂ by dividing the VCO output by N as shownin FIG. 31( b). The output of the VCO 864 (at f₁) is coupled to adivider 872. The output of te divider 872 (at f₂) is mixed with the VCOat mixer 868 to produce an RF clock frequency equal to:f_(LO)=f₁′(1+1/N), where f₁ is the VCO frequency. A bandpass filter 874at the mixer output can be used to reject the lower sideband located atf₁-f₁/N.

In another embodiment of the present invention, a single sideband mixingscheme is used for the LO generator. FIG. 32 shows a single sidebandmixing scheme. This approach generates I and Q signals at the VCO 864output. The output of the VCO 864 is coupled to a quadrature frequencydivider 876 should be able to deliver quadrature outputs. Quadratureoutputs will be realized if the divide ratio (N) is equal to two to thepower of an integer (N=2^(n)). The I signal output of the divider 876 ismixed with the I signal output of the VCO 864 by a mixer 878. Similarly,the Q signal output of the divider 876 is mixed with the Q signal outputof the VCO 864 by a mixer 880.

Although a single sideband structure uses two mixers, this should notdouble the mixer power consumption, since the gain of the singlesideband mixer will be twice as much. By utilizing a Gilbert cell (i.e.,a current commutating mixer) for each mixer 878, 880, the addition orsubtraction required in a single sideband mixer can be done byconnecting the two mixers 878, 880 outputs and sharing a common load(e.g., an LC circuit). The current from the mixers is added orsubtracted, depending on the polarity of the inputs, and then convertedto a voltage by an LC load (not shown) resonating at the desiredfrequency.

FIG. 33 shows an LO generator architecture in accordance with anembodiment of the present invention. This architecture is similar to thearchitecture shown in FIG. 32, except that the LO generator architecturein FIG. 33 generates I-Q data. In a low-IF system, a quadrature LO isdesirable for image rejection. In the described embodiment, the I and Qoutputs of the VCO can be applied to a pair of single sideband mixer togenerate quadrature LO signals. A quadrature VCO 48 produces I and Qsignals at its output. Buffers are included to provide isolation betweenthe VCO output and the LO generator output. The buffer 884 buffers the Ioutput of the VCO 48. The buffer 886 buffers the Q output of the VCO 48.The buffer 888 combines the I and Q outputs of the buffers 884, 886. Thesignal from the buffer 888 is coupled to a frequency divider 890 whereit is divided by N and separated into I and Q signals. The I-Q outputsof the divider 890 are buffered by buffer 892 and buffer 894. The Ioutput of the divider 890 is coupled to a buffer 892 and the Q signaloutput of the divider 890 is coupled to a buffer 894. A first mixer 896mixes the I signal output of the buffer 892 with the I signal output ofthe buffer 884. A second mixer 897 mixes the Q signal output from thebuffer 894 with Q signal output from the buffer 886. A third mixer 898mixes the Q signal output of the buffer 894 with the I signal output ofthe buffer 884. A fourth mixer 899 mixes the I signal output from thebuffer 892 with the Q signal output from the buffer 886. The outputs ofthe first and second mixers 896, 897 are combined and coupled to buffer900. The outputs of the third and fourth mixers 898, 899 are combinedand coupled to buffer 902. LC circuits (not shown) can be positioned atthe output of each buffer 900, 902 to provide a second-order filterwhich rejects the spurs and harmonics produced due to the mixing actionin the LO generator.

Embodiments of the present invention which are integrated into a singleIC may employ buffers configured as differential pairs with a currentsource to set the bias. With this configuration, if the amplitude of thebuffer input is large enough, the signal amplitude at the output will berather independent of the process parameters. This reduces thesensitivity of the design to temperature or process variation.

The lower sideband signal is ideally rejected with the describedembodiment of the LO generator because of the quadrature mixing.However, in practice, because of the phase and amplitude inaccuracy atthe VCO and divider outputs, a finite rejection is obtained. In singleIC fully integrated embodiments of the present invention, the rejectionis mainly limited to the matching between the devices on chip, and istypically about 30-40 dB. Since the lower sideband signal is 2×f₁/N awayin frequency from the desired signal, by proper choice of N, it can befurther attenuated with on-chip filtering.

Because of the hard switching action of the buffers, the mixers willeffectively be switched by a square-wave signal. Thus, the divideroutput will be upconverted by the main harmonic of VCO (f₁), as well asits odd harmonics (n×f₁), with a conversion gain of 1/n. In addition, atthe input of the mixer, because of the nonlinearity of the mixers, andthe buffers preceding the mixers, all the odd harmonics of the inputsignals to the mixers will exist. Even harmonics, both at the LO and theinput of the mixers can be neglected if a fully balanced configurationis used. Therefore, all the harmonics of VCO (n×f₁) will mix with allthe harmonics of input (m×f₂), where f₂ is equal to f₁/N. Because of thequadrature mixing, at each upconversion only one sideband appears at themixer output. Upper or lower sideband rejection depends on the phase ofthe input and LO at each harmonic. For instance, for the main harmonicsmixed with each other, the lower sideband is rejected, whereas when themain harmonic of the VCO mixes with the third harmonic of the divideroutput signal, the upper sideband is rejected. Table 1 gives a summaryof the cross-modulation products up to the 5^(th) harmonic of the VCOand input. In each product, only one sideband is considered, since theother one is attenuated due to quadrature mixing, and is negligible.

All the spurs are at least 2×f₁/N away from the main signal located atf₁×(1+1/N). The VCO frequency will be f₁/N away from the PA output.Thus, by choosing a smaller N better filtering can be obtained. Inaddition, the VCO frequency will be further away from the PA outputfrequency. The value of N, and the quality factor (Q) of the resonators(not shown) positioned at the output of each component determine howmuch each spur will be attenuated. The resonator quality factor isusually set by the inductor Q, and that depends merely on the ICtechnology. Higher Q provides better filtering and lower powerconsumption.

TABLE 1 Cross-Modulation Products at the LO Generator Output 1^(st):f₁/N 3^(rd): 3f₁/N 5^(th): 5f₁/N 1^(st): f₁ f₁ × (1 + 1/N) f₁ × (1 −3/N) f₁ × (1 + 5/N) 3^(rd): 3f₁ f₁ × (3 − 1/N) f₁ × (3 + 3/N) f₁ × (3 −5/N) 5^(th): 5f₁ f₁ × (5 + 1/N) f₁ × (5 − 3/N) f₁ × (5 + 5/N)

The maximum filtering is obtained by choosing N=1. Moreover, in thiscase, the frequency divider is eliminated. This lowers the powerconsumption and reduces the system complexity of the LO generator.However, the choice of N=1 may not be practical for certain embodimentsof the present invention employing a low-IF receiver architecture withquadrature LO signals. The problem arises from the fact that the thirdharmonic of the VCO (at 3 f₁) mixed with the divider output (at f₁) alsoproduces a signal at 2 f₁ which has the same frequency as the maincomponent of the RF clock output from the LO generator. With theconfiguration shown in FIG. 33, the following relations hold for themain harmonics:Cos(ω₁ t)·Cos(ω₁ t)−Sin(ω₁ t)·Sin(ω₁ t)→Cos(2ω₁ t)  (45)andCos(ω₁ t)·Sin(ω₁ t)+Sin(ω₁ t)·Cos(ω₁ t)→Sin(2ω₁ t)  (46)which show that at the output of the mixers, quadrature signals at twicethe VCO frequency exist. For the VCO third harmonic mixed with thedivider output, however, the following relations hold:−Cos(ω₁ t)·⅓ Cos(3ω₁ t)−Sin(ω₁ t)·⅓ Sin(3ω₁ t)→−⅓ Cos(2ω₁ t)  (47)andCos(ω₁ t)·⅓ Sin(3ω₁ t)−Sin(ω₁ t)·⅓ Cos(3ω₁ t)→−⅓ Sin(2ω₁ t)  (48)The factor ⅓ appears in the above equations because the third harmonicof a square-wave has an amplitude which is one third of the mainharmonic. Comparing equation (46) with equation (48), the two productsare added in equation (46), while they are subtracted in equation (47).The reason is that for the main harmonic of the VCO, quadrature outputshave phases of 0 and 90°, whereas for the third harmonic, the phases are0 and 270°. The same holds true for equation (45) and equation (47). Thetwo cosines in equation (45) and equation (47), when added, give acosine at 2ω₁ with an amplitude of ⅔, yet the two sinewaves in equation(46) and equation (48) when added, give a component at 2ω₁ with anamplitude of 4/3. Therefore, a significant amplitude imbalance exists atthe I and Q outputs of the mixers. When these signals pass through thenonlinear buffer at the mixers output, the amplitude imbalance will bereduced. However, because of the AM to PM conversion, some phaseinaccuracy will be introduced. The accuracy can be improved with aquadrature generator, such as a polyphase filter, after the mixers. Apolyphase filter, however, is lossy, especially at high frequency, andit can load its previous stage considerably. This increases the LOgenerator power consumption significantly, and renders the choice of N=1unattractive for embodiments of the present invention employing a low-IFreceiver architecture with quadrature LO signals.

For N=2, the LO generator output will have a frequency of 1.5 f₁ and theclosest spurs will be located ±f₁ away from the output. These spurs canbe rejected by positioning LC filters (not shown) at the output of eachcircuit in the LO generator. A second-order LC filter tuned to f₀, witha quality factor Q, rejects a signal at a frequency of f as given in thefollowing equation:

$\begin{matrix}{{{H(f)}} = \frac{\frac{f}{{Qf}_{0}}}{\sqrt{\left\lbrack {1 - \left( \frac{f}{f_{0}} \right)^{2}} \right\rbrack^{2} + \left( \frac{f}{{Qf}_{0}} \right)^{2}}}} & (49)\end{matrix}$

The following discussion changes based on the Q value. Considering a Qof about 5 for the inductor, with f₀=1.5 f₁, the spur located at 2.5 f₁is rejected by about 15 dB by each LC circuit. This spur is produced atthe LO generator output due to the mixing of the VCO third harmonic (at3 f₁) with the divider output (at 0.5 f₁). This signal is attenuated by10 dB since the third harmonic of a square-wave is one third of the mainharmonic, 15 dB at the LC resonator at the mixers output tuned to 1.5f₁, and another 15 dB at the output of the buffers (900, 902 in FIG.33). This gives a total rejection of 40 dB. When applied to the mixersin the transmitter, this LO generator output will upconvert the basebanddata to 2.5 f₁. With LC filters (not shown) positioned at theupconversion mixers and PA output in the transmitter, another 15+15=30dB rejection is obtained (FIG. 33).

The spur located at 0.5 f₁ is produced because of the third harmonic ofthe divider output (at 1.5 f₁) is mixed with the VCO output (at f₁).Because of the hard switching action at the divider output, the thirdharmonic is about 10 dB lower than the main harmonic at 0.5 f₁. Thebuffer at the divider output tuned to 0.5 f₁ (892, 8943 in FIG. 33),rejects this signal by about 22 dB (equation (24)). This spur can befurther attenuated by LC circuits at the mixer and its buffer output by(2)(22)=44 dB. The total rejection is 76 dB.

FIG. 33( a) shows a signal passing through a limiting buffer 910 (suchas the buffers implemented in the LO generator). When a large signal ata frequency of f accompanied with a small interferer at a frequency ofΔf 902 away pass through a limiting buffer, at the limiter output theinterferer produces two tones ±Δf 914, 916 away from the main signal,each with 6 dB lower amplitude. Therefore, the spur at 2.5 f₁ willactually be 10+15+15+6=46 dB attenuated when it passes through thebuffer, instead of the 40 dB calculated above. It will also produce animage at 0.5 f₁ which is 10+15+22+6=53 dB lower than the main signal.This will dominate the spur at 0.5 f₁ because of the third harmonic ofthe divider mixed with the VCO signal, which is more than 75 dB lowerthan the main signal.

Since the buffer is nonlinear, another major spur at the LO generatoroutput is the third harmonic of the main signal located at 3×1.5 f₁.This signal will be 10+22=32 dB lower than the main harmonic. The 22 dBrejection results from an LC circuit (not shown) tuned to 1155 f₁(equation (49)) in the buffer. This undesired signal will not degradethe LO generator performance, since even if a perfect sinewave isapplied to upconversion (or downconversion) mixers, due to hardswitching action of the buffer, the mixer is actually switched by asquare-wave whose third harmonic is only 10 dB lower. Thus, if anonlinear PA is used in the transmitter, even with a perfect input tothe PA, the third harmonic at the transmitter output will be 10+22+10=42dB lower. The first 10 dB is because the third harmonic of a square-waveis one third of the main one, the 22 dB is due to the LC filter at thePA output, and the last 10 dB is because the data is spread in thefrequency domain by three times. Any DC offset at the mixer input in thetransmitter is upconverted by the LO, and produces a spur at f₁. Thisspur can be attenuated by 13 dB for each LC circuit used (equation(49)). In addition, the signal at the mixer input in the transmitter isconsiderably larger (about 10-20 times) than the DC offset. Thus thespur at f; will be about 13+13+26=52 dB lower than the main signal. Allother spurs given in Table 1 are more than 55 dB lower at the LOgenerator output. The dominant spur is the one at 2.5 f₁ which is about46 dB lower than the main signal.

Choosing N>2 may not provide much benefit for single IC embodiments ofthe present invention with the possible exception that the on-chipfiltering requirements may be relaxed. When using an odd number for N,further disadvantages may be realized because the divider output willnot be in quadrature thereby preventing single sideband mixing. Inaddition, for N>2 the divider becomes more complex and the powerconsumption increases. Nevertheless, in certain applications, N=4 may beselected over N=2 so that the divider quadrature accuracy will notdepend on the duty cycle of the input signal.

When choosing N equal to 2^(n), such as N=2, quadrature signals arereadily available at the divider output despite quadrature phaseinaccuracies at the output of the VCO. Assume that the VCO outputs havephase of 0 and 90°+q, where q is ideally 0, and that the dividerproduces perfect quadrature outputs. At the LO generator outputs thefollowing signals exist:V _(out) _(—) _(I)=Cos(ω₂ t)·Cos(ω₁ t+θ)−Sin(ω ₂ t)·Sin(ω₁ t)  (50)andV _(out) _(—) _(Q)=Cos(ω₂ t)·Sin(ω₁ t)+Sin(ω₂ t)·Cos(ω₁ t+θ)  (51)where ω₁ is the VCO radian frequency, and ω₂ is the divider radianfrequency, equal to 0.5ω₁. By simplifying equation (25) and equation(26), the signals at the output of mixers will be:

$\begin{matrix}{{V_{out\_ I} = {{{- {{Sin}\left( \frac{\theta}{2} \right)}} \cdot {{Sin}\left( {{\left( {\omega_{1} - \omega_{2}} \right)t} + \frac{\theta}{2}} \right)}} + {{{Cos}\left( \frac{\theta}{2} \right)} \cdot {{Cos}\left( {{\left( {\omega_{1} + \omega_{2}} \right)t} + \frac{\theta}{2}} \right)}}}}{and}} & (52) \\{V_{out\_ Q} = {{{- {{Sin}\left( \frac{\theta}{2} \right)}} \cdot {{Cos}\left( {{\left( {\omega_{1} - \omega_{2}} \right)t} + \frac{\theta}{2}} \right)}} + {{{Cos}\left( \frac{\theta}{2} \right)} \cdot {{Sin}\left( {{\left( {\omega_{1} + \omega_{2}} \right)t} + \frac{\theta}{2}} \right)}}}} & (53)\end{matrix}$

The above equations show that regardless of the value of θ, the outputsare always in quadrature. However, other effects should be evaluated.First, a spur at ω₁−ω₂=0.5ω₁ is produced at the output. This spur can beattenuated by 2×22=44 dB by the LC filters at the mixer and its bufferoutputs. Thus, for 60 dB rejection, the single sideband mixers need toprovide an additional 16 dB of rejection (about 0.158). Based onequation (53), tan(θ/2)=0.158, or θ≈18°, phase accuracy of better than18° can generally be achieved. Second, phase error at the VCO outputlowers the mixer gain (term Cos(θ/2) in equation (52) or (53)). For aphase error of 18°, the gain reduction is, however, only 0.1 dB, whichis negligible. For θ=90° (a single-phase VCO), both sidebands areequally upconverted at the mixer output. However, the LC filters rejectthe lower sideband by about 44 dB. The mixer gain will also be 3 dBlower. This will slightly increase the power consumption of the LOgenerator. If θ=180° (the VCO I and Q outputs are switched), the lowersideband is selected, and the desired sideband is completely rejected.

Similarly, the LO generator will not be sensitive to the phase imbalanceof the divider outputs if the VCO is ideal. However, if there is somephase inaccuracy at both the divider and VCO outputs, the LO generatoroutputs will no longer be in quadrature. In fact, if the VCO output hasa phase error of q₁ and the divider output has a phase error of q₂, theLO generator outputs will be:

$\begin{matrix}{{V_{out\_ I} = {{{- {{Sin}\left( \frac{\theta_{1} - \theta_{2}}{2} \right)}} \cdot {{Sin}\left( {{\left( {\omega_{1} - \omega_{2}} \right)t} + \frac{\theta_{1} - \theta_{2}}{2}} \right)}} + {{{Cos}\left( \frac{\theta_{1} + \theta_{2}}{2} \right)} \cdot {{Cos}\left( {{\left( {\omega_{1} + \omega_{2}} \right)t} + \frac{\theta_{1} + \theta_{2}}{2}} \right)}}}}{and}} & (54) \\{V_{out\_ Q} = {{{- {{Sin}\left( \frac{\theta_{1} + \theta_{2}}{2} \right)}} \cdot {{Cos}\left( {{\left( {\omega_{1} - \omega_{2}} \right)t} + \frac{\theta_{1} - \theta_{2}}{2}} \right)}} + {{{Cos}\left( \frac{\theta_{1} - \theta_{2}}{2} \right)} \cdot {{Sin}\left( {{\left( {\omega_{1} + \omega_{2}} \right)t} + \frac{\theta_{1} + \theta_{2}}{2}} \right)}}}} & (55)\end{matrix}$

This shows that the outputs still have phases of 0 and 90°, but theiramplitudes are not equal. The amplitude imbalance is equal to:

$\begin{matrix}{\frac{\Delta\; A}{A} = {{2\frac{{{Cos}\left( \frac{\theta_{1} + \theta_{2}}{2} \right)} - {{Cos}\left( \frac{\theta_{1} - \theta_{2}}{2} \right)}}{{{Cos}\left( \frac{\theta_{1} + \theta_{2}}{2} \right)} + {{Cos}\left( \frac{\theta_{1} - \theta_{2}}{2} \right)}}} = {2{\tan\left( \frac{\theta_{1}}{2} \right)} \times {\tan\left( \frac{\theta_{2}}{2} \right)}}}} & (56)\end{matrix}$

If θ₁ and θ₂ are small and have an equal standard deviation, that is,the phase errors in the VCO and divider are the same in nature, then theoutput amplitude standard deviation will be:

$\begin{matrix}{\sigma_{A} \approx \frac{\left( \sigma_{\theta} \right)^{2}}{2}} & (57)\end{matrix}$where σ_(A) is the standard deviation of the output amplitude, and σ_(θ)is the phase standard deviation in radians. Equation (57) denotes thatthe phase inaccuracy in the VCO and divider has a second order effect onthe LO generator. For instance, if θ₁ and θ₂ are on the same order andabout 10°, the amplitude imbalance of the output signals will be onlyabout 1.5%. In this case, the lower sideband will be about 15 dBrejected by the mixers, which will lead to a total attenuation of about22+22+15=59 dB. This shows that the LO generator is robust to phaseerrors at the VCO or divider outputs, since typically phase errors ofless than 5° can be obtained on chip.

Phase errors in the divider can originate from the mismatch at itsoutput. Moreover, for N=2, if the input of the divider does not have a50% duty cycle, the outputs will not be in quadrature. Again, thedeviation from a 50% duty cycle in the divider input signal may becaused due to mismatch. Typically, with a careful layout, this mismatchis minimized to a few percent. The latter problem can also be alleviatedby improving the common-mode rejection of the buffer preceding thedivider (888 in FIG. 33). One possible way of doing so is to add a smallresistor at the common tail of the inductors in the buffer. For adifferential output, this resistor does not load the resonator at thebuffer output, since the inductors common tail is at AC ground. Acommon-mode signal at the output is suppressed however, since thisresistor degrades the LC circuit quality factor. The value of theresistor should be chosen appropriately so as not to produce a headroomproblem in the buffer.

Embodiments of the present invention that are fully integrated onto asingle IC can be implemented with a wide tuning range VCO with constantgain. In a typical IC process, the capacitance can vary by 20%. Thistranslates to a 10% variation in the center frequency of the oscillator.A wide tuning range can be used to compensate for variation. Variationsin temperature and supply voltage can also shift the center frequency.To generate a wide tuning range, two identical oscillators can becoupled together as shown in FIG. 34. This approach forces theoscillation to be dependent on the amount of coupling between the twooscillators.

In the described exemplary embodiment of the VCO shown in FIG. 34, thetuning curve is divided into segments with each segment digitallyselected. This approach ensures a sufficient amount of coupling betweenthe two oscillators for injection lock. In addition, good phase noiseperformance is also obtained. The narrow frequency segment prevents thegain of the VCO from saturating. The segmentation lowers the VCO gain bythe number of segments, and finally by scaling the individual segments,a piecewise linear version of the tuning curve is made resulting in aconstant gain VCO.

FIG. 34 shows a block diagram of the wide tuning range VCO comprisingtwo coupled oscillators where the amount of coupling transconductance isvariable. The wide tuning range VCO comprises two resonators 800, 802and four transconductance cells, g_(m) cells 804, 806, 808, 810. Thetransconductance cells are driver that converts voltage to current. Thetransconductance cells used to couple the oscillators together have avariable gain. The first VCO 800 provides the I signal and the secondVCO provides the Q signal. The output of the first VCO 800 and theoutput of the second VCO 802 are coupled to transconductance cells 806,807, respectively, combined, and fed back to the first VCO 800. Thetransconductance cell 807 used for feeding back the output of the secondVCO to the first VCO is a programable variable gain cell. Similarly, theoutput of the second VCO 802 and the output of the first VCO 800 arecoupled to transconductance cells 805, 804, respectively, combined, andfed back to the second VCO 802. The transconductance cell 804 used forfeeding back the output of the first VCO to the second VCO is aprogrammable variable gain cell. The gain of the programmable variablegain transconductance cells 804, 807 can be digitally controlled fromthe controller.

FIG. 35 shows a schematic block diagram of the wide-tuning range VCOdescribed in connection with FIG. 34. The wide-tuning range VCO includesindividual current sources 810, 812, 814, 816, cross-coupled transistors818, 820 with resonating inductors 826, 828, and cross-coupledtransistors 822, 824 with resonating inductors 830, 832. Twodifferential pairs couple the two sets of oscillators. Differential pair834, 836 are coupled to the drains of transistors 824, 822,respectively. Differential pair 838, 840 are coupled to the drains oftransistors 818, 820. Tank #1 comprises inductors 826 and 828. Tank #2comprises inductors 830 and 832.

Transistors 818 and 820 form a cross-coupled pair that injects a currentinto tank #1 in which the current through the transistor 818 is exactly180 degrees out of phase with the current in the transistor 820.Likewise, transistors 822 and 824 form a cross-coupled pair that injectsa current into tank #2 in which the current through the transistor 822is exactly 180 degrees out of phase with the current in the transistor824. The first set of coupling devices 834, 836 injects a current intotank #1 that is 90 degrees out of phase with current injectedrespectively by the transistors 818, 820. The second set of couplingdevices 838, 840 injects a current into tank #2 that is 90 degrees outof phase with the current injected respectively by the transistors 822,824. The tank impedances causes a frequency dependent phase shift. Byvarying the amplitude of the coupled signals, the frequency ofoscillation changes until the phase shift through the tanks results in asteady-state solution. Varying the bias of the current source controlsthe gm of the coupling devices. Current sources 812, 816 provide controlof VCO tuning. Current sources 810, 814 provide segmentation of the VCOtuning range.

FIG. 36( a) shows the typical tuning curve of the wide tuning range VCObefore and after segmentation. The horizontal axis is voltage. Thevertical axis is frequency. FIG. 36( b) shows how segmentation is usedto divide the tuning range and linearize the tuning curve. The lineartuning curves correspond to different VCO segments. The slope of thelinear tuning curves is a result control of VCO tuning. The horizontalaxis is voltage. The vertical axis is frequency.

FIG. 37( a) shows how the VCO of FIG. 34 can be connected to the dividerbefore being upconverted to the RF clock frequency in the LO generator.The I output signal of the VCO is coupled to buffer 884 and the Q outputsignal of the VCO is coupled to buffer 886. Buffer 888 combines the I-Qdata from the buffer 884 and the buffer 886 to obtain a larger signal.The large signal is coupled to a divider 50 where it is divided infrequency by N to get quadrature signals.

In another embodiment of the present invention, a polyphase filter 892follows a single-phase VCO as shown in FIG. 37( b). This approach uses asingle phase VCO 48 with a polyphase filter 892 to get quadraturesignals. The output of the VCO 48 is coupled to a buffer 888. The bufferprovides sufficient drive for the polyphase filter 892.

A multiple stage polyphase filter can be used to obtain better phaseaccuracy at a certain frequency range. Embodiments of the presentinvention that are fully integrated into a single IC, the requiredfrequency range is mainly set by the process variation on the chip andthe system bandwidth.

Any amplitude imbalance in the signals at the VCO and divider outputwill only cause a second order mismatch in the amplitude of the LOgenerator signals, and the output phase will remain 0 and 90°. If thestandard deviation of the amplitude imbalance at the VCO and divider arethe same and equal to σ_(a), then the standard deviation of the LOgenerator output amplitude imbalance (σ_(A)) will be:

$\begin{matrix}{\sigma_{A} = \frac{\left( \sigma_{\alpha} \right)^{2}}{2}} & (58)\end{matrix}$

The reason phase inaccuracy is more emphasized here is that because ofthe limiting stages in the LO generator and the hard switching at themixers LO input, most of the errors will be in phase, rather thanamplitude.

Although the phase or amplitude inaccuracy at the mixers input or LO hasonly a second order effect on the LO generator, any mismatch at themixers outputs or the following stages will directly cause phase andamplitude imbalance in the LO generator outputs. This mismatch willtypically be a few percent, and will not adversely impact thetransceiver performance, since in a low-IF or direct conversionarchitectures the required image rejection is usually relaxed.

4.0 Controller

The controller performs adaptive programming and calibration of thereceiver, transmitter and LO generator (see FIG. 2). An exemplaryembodiment of the controller in accordance with one aspect of thepresent invention is shown in FIG. 38. A control bus 17 provides two waycommunication between the controller and the external processing device(not shown). This communication link can be used to externally programthe transceiver parameters for different modulation schemes, data ratesand IF operating frequencies. In the described exemplary embodiment, theexternal processing device transmits data across the control bus 17 to abank of addressable registers 900-908 in the controller. Eachaddressable register 900-908 is configured to latch data for programmingone of the components in the transmitter, receiver LO generator. By wayof example, the power amplifier register 900 is used to program the gainof the power amplifier 62 in the transmitter (see FIG. 2). The LOregister 902 is used to program the IF frequency in the LO generator.The demodulator register 903 is used to program the demodulator for FSKdemodulation, or alternatively in the described exemplary embodiment,program the A/D converter to handle different modulation schemes. TheAGC register 905 programs the gain of the programmable multiple stageamplifier when in the AGC mode. The filter registers 901, 904, 906program the frequency and bandwidth of their respective filters.

The transmission of data between the external processing device and thecontroller can take on various forms including, by way of example, aserial data stream parsed into a number of data packets. Each datapacket includes programming data for one of the transceiver componentsaccompanied by a register address. Each register 900-908 in thecontroller is assigned a different address and is configured to latchthe programming data in the each data packet where the register addressin that data packet matches its assigned address.

The controller also may include various calibration circuits. In thedescribed exemplary embodiment, the controller is equipped with an RCcalibration circuit 907 and a bandgap calibration circuit 908. The RCcalibration circuit 907 can compensate an integrated circuit transceiverfor process, temperature, and power supply variations. The bandgapcalibration circuit can be used by the receiver, transmitter, and LOgenerator to set amplifier gains and voltage swings.

The programming data from the addressable registers 900-908 and thecalibration data from the RC calibration circuit 907 and the bandgapcalibration circuit 908 are coupled to an output register 909. Theoutput register 909 formats the programmability and calibration datainto a data packets. Each data packet includes a header or preamblewhich addresses the appropriate transceiver component. The data packetsare then transmitted serially over a controller bus 910 to their finaldestination. By way of ex ample, the output register 909 packages theprogramming data from the power amplifier register 900 with the headeror preamble for the power amplifier and outputs the packaged data as thefirst data packet to the controller bus 910.

The second data packet generated by the output register 909 is for theprogrammable low pass filter in the transmitter. The second data paketincludes two data segments each with its own header or preamble. Thefirst segment consists of both programmability and calibration data.Because the programmability feature requires a large dynamic range asfar as programming the programmable low pass filter to handle differentfrequency bands, and the calibration feature is more of a fine tuningfunction of the programmable low pass filter once tuned requiring a muchsmaller dynamic range, a single digital word containing both programmingand calibration information can be used with the most significant bits(MSB) having the programming information and the least significant bits(LSB) having the calibration information. To this end, the outputregister 909 combines the output of the low pass filter register 901with the output of the RC calibration circuit 907 with the low passfilter register output constituting the MSBs and the RC calibrationcircuit output constituting the LSBs. A header or preamble is attachedto the combined outputs identifying the data packet for RC calibrationof the programmable low pass filter in the transmitter. Similarly, thesecond segment of the second data packet is generated by combining thelow pass filter register output (as the MSBs) with the bandgapcalibration circuit output (as the LSBs) and attaching a header orpreamble identifying the data packets for bandgap calibration of theprogrammable low pas filter.

The third data packet generated and transmitted by the output register909 can program the dividers in the LO generator to produce different IFfrequencies. The third data packet can be a single segment of data witha header or preamble identifying the LO generator for programming eachdivider. Alternatively, the third data packet can include any number ofdata segments with, in one embodiment, different programming data foreach divider in the LO generator. Each data segment would include aheader or preamble identifying a specific divider in the LO generator.

The fourth data packet generated and transmitted by the output register909 could include the programming data output from the demodulatorregister 904 with the appropriate header or preamble.

The output of the complex bandpass filter register 904 can be combinedwith the output from the RC calibration circuit 907 to form the firstsegment of the fifth data packet. The output of the complex bandpassfilter register 904 can also be combined with the output of the bandgapcalibration circuit 908 to form the second segment of the fifth datapacket. Each segment can have its own header or preamble indicating thetype of calibration data for the complex bandpass filter.

The sixth data packet generated and transmitted by the output register909 can be the output data from the AGC register 905 accompanied by aheader or preamble identifying the data packet for the programmablemultiple stage amplifier in the receiver.

The output of the polyphase filter register 906 can be combined with theoutput from the RC calibration circuit 907 to form the first segment ofthe seventh data packet. The output of the polyphase filter register 906can also be combined with the output of the bandgap calibration circuit908 to form the second segment of the seventh data packet. Each segmentcan have its own header or preamble indicating the type of calibrationdata for the polyphase filter.

Finally, the output register 907 can configure additional data packetsfrom the output of the RC calibration circuit 907 and, in separate datapackets, the output of the bandgap calibration circuit 908 withappropriate headers or preambles.

As those skilled in the art will appreciate, other data transmissionschemes can be used. By way of example, the separate output registersfor each transceiver component could be used. In this embodiment, eachoutput register would be directly connected to one or more transceivercomponents.

4.1 RC Calibration Circuit

RC calibration circuits can provide increased accuracy for improvedperformance. Embodiments of the present invention that are integratedinto a single IC can utilize RC calibration to compensate for process,temperature, and power supply variation. For example, variations in theabsolute value of the RC circuit in a complex filter can limit theamount of rejection that the filter can provide. In the describedexemplary embodiments of the present invention, an RC calibrationcircuit in the controller can provide dynamic calibration of every RCcircuit by providing a control word to the transmitter, receiver and LOgenerator.

FIG. 39 shows an exemplary RC calibration circuit in accordance with anembodiment of the present invention. The calibration circuit uses thereference clock from the LO generator to generate a 4-bit control wordusing a compare-and-increment loop until an optimum value is obtained.The 4-bit control provides an efficient technique for calibrating the RCcircuits of the transceiver with a maximum deviation from its optimalvalue of only 5%.

Transistors 172, 174, 176. 178, 180, 182 form a cascode current sourcewith a reference current I_(REF) 184. With the gates of the transistors172 and 178 tied to their respective sources, a fixed reference currentI_(REF) 184 can be established. By tying the gates of the transistors174, 180 to the gates of the transistors 172, 178, respectively, thecurrent through resistor R_(C) 186 can be mirrored to I_(REF) 184.Similarly, by tying the gates of the transistors 176, 182 to the gatesof the transistors 174, 180, respectively, the current through resistorR_(C) 186 can be mirrored to a tunable capacitor C_(C) 188. Thecalibration circuit tunes the absolute value of the RC to a desiredfrequency by using this cascode-current source to provide identicalcurrents to the on-chip reference resistor R_(C) 186 and to the tunablecapacitor C_(C) 188 generating the voltages V_(RES) 190 and V_(CAP) 192,respectively. Embodiments of the present invention that are integratedinto a single IC can use an off-chip reference resistor R_(c) to obtaingreater calibration accuracy. The current through the tunable capacitoris controlled by a logic control block 195 via switch S₂ 193. During thecharging phase, switch S₂ 193 is closed and switch S₁ is open to chargethe tunable capacitor C_(C) 188 to V_(CAP). The voltage held on thetunable capacitor 188 V_(CAP) is then compared, using a latchedcomparator 198, to a voltage generated across the reference resistor186. The value of the tunable capacitor C_(c) 188 is incremented insuccessive steps by the logic control block 195 until the voltage heldby the tunable capacitor C_(C) matches the voltage across the referenceresistor 186, at which point the 4-bit control word for optimalcalibration of the RC circuits for the transmitter, receiver, and LOgenerator is obtained. More particularly, once the voltage V_(CAP)reaches the voltage V_(RES), the output of the comparator output 198switches. The switched comparator output is detected by the controllogic 195. The control logic 195 opens switch S₂ 193 and closes switchS₁ 194 causing the tunable capacitor 188 C_(C) to discharge. Theresultant 4-bit control word is latched by the control logic 195 andcoupled to the transceiver, receiver, and LO generator.

C_(p) 200 compensates for the parasitic capacitance loading of thecapacitive branch. By choosing C_(c) 188 to be much larger than C_(p)200, the voltage error at node V_(CAP) 192 caused by, charging theparasitic capacitance becomes negligible.

The clock signals used by the calibration circuit are generated by firstdividing the reference clock down in frequency, and then converting theresult into different phases for the charging, comparison, increment,and discharging phases of calibration. Embodiments of the presentinvention that are integrated onto a single IC can obtain an accurate RCvalue because capacitor scaling and matching on the same integratedcircuit can be well-controlled with proper layout technique. Thedescribed RC calibration circuit provides an RC-tuning range ofapproximately +40%, which is sufficient to cover the range of processvariation typical in semiconductor fabrication.

4.2 RC Calibration Circuit Using Polyphase Filtering

An RC calibration circuit using polyphase filtering is an alternativemethod for calibrating RC circuits in the transmitter, receiver, and LOgenerator. The RC calibration using polyphase filtering circuit includesan auto-calibration algorithm in which the capacitors or the RC circuitsin the transceiver, receiver and LO generator can be calibrated with acontrol word generated by comparing the signal attenuation across twotunable polyphase filters. The calibrated RC value obtained as a resultof this algorithm is accurate to within ±5% of its optimal value.

FIG. 40 shows an exemplary embodiment of the RC calibration circuitusing polyphase filtering. The RC calibration circuit uses the referenceclock from the LO generator to adjust the RC value in two polyphasefilters 280, 282 in successive steps until an optimum value has beenselected. In this process, the two polyphase filters 280, 282 providesignal rejection that is dependent upon the value of ω=(RC)⁻¹ to whichthey are tuned by control logic 286. Initially, the first filter(Polyphase A) 280 is tuned to a frequency less than the frequency of thereference clock (reference frequency), and the second filter (PolyphaseB) 282 is tuned to a frequency greater than the reference frequency bycontrol logic 286. The signals at the outputs of the polyphase filtersare detected with a received-signal-strength-indicator (RSSI) block 284,285 in each path. A filter is coupled to RSSI block 284 and thepolyphase B filter is coupled to RSSI block 285.

With an input dynamic range of 50 dB, the RSSI circuit is designed todetect the levels of rejection provided by the polyphase filtering. Theoutputs of RSSI block 284 and RSSI block 285 are coupled to a comparator280 where the level of signal rejection of each polyphase filter iscompared by comparator 280. The outputs of the RSSI blocks are alsocoupled to the control logic 286. The control logic 286 determines fromthe RSSI outputs which polyphase filter has a lower amount of signalsuppression. Then, the control logic 286 adjusts the frequency tuning ofthat filter in an incremental step via the control logic 286. This isdone by either increasing the tuned frequency of the first filter(polyphase A) filter 280, or by decreasing the tuned frequency of thesecond filter (polyphase B) 282 by changing the appropriate 4-bitcontrol word. This process continues in successive steps until the 4-bitcontrol word in each branch are identical, at which point, the RC valuesof the two polyphase filters are equal. The 4-bit control word providesa maximum deviation of only ±5%.

In the described exemplary embodiment, the frequency of the input signalX_(IN) is derived from the reference frequency and is chosen to be, byway of example, 2 MHz. This input signal X_(IN) is obtained by initiallydividing the reference clock down in frequency, followed by a conversioninto quadrature phases at the control logic 286. By dividing thereference clock by a factor greater than two with digital flip-flops(not shown), the input signal at X_(IN) is known to be differential withwell-defined quadrature phases.

Two branches of polyphase filtering are used in this algorithm. Two4-bit control words are used to control the value of the capacitances ineach polyphase filter. The initial control words set the capacitance inthe first filter (Polyphase A) to its maximum value and the capacitancein the second filter (Polyphase B) to its minimum value. This providesan initial condition in which the filters have maximum signalsuppression set at frequencies (ω_(low) and ω_(high)) that areapproximately ±40% of the frequency of the input signal X_(IN) for thecase of nominal process variation. For a sinusoidal input X_(IN), thecalibration circuit depicted in FIG. 40 would require only asingle-stage polyphase filter in each branch The single-stage filterswould attenuate the sinusoid input signal, generating outputs at X_(A)and X_(B) with the dominant one still at the same frequency as the inputsignal. However, the reference clock from the LO generator is a digitalrail-to-rail clock. Because the input is not a pure sinusoid,multiple-stage filters may provide greater calibration accuracy. In thecase of a single-stage filter with a digital clock, the filter wouldsuppress the fundamental frequency component at ω_(in) to a significantdegree but the harmonics would pass through relatively unaffected. TheRSSI block would then detect and limit the third harmonic component ofthe input signal at 3ω_(low), as it becomes the dominant frequencycomponent after the fundamental is suppressed. This could result in aninaccurate calibration code.

A three-stage polyphase filter can be used in each branch to suppressthe fundamental frequency component of X_(IN) as well as the 3rd and 5thharmonics. The first stage of the polyphase filter can provide rejectionof the fundamental frequency component The second stage can providerejection of the 3rd harmonic. The third stage can provide rejection ofthe 5th harmonic. At the same time, the higher harmonics of the inputsignal X_(IN) can be suppressed with an RC lowpass filter in a buffer(not shown) preceding the polyphase filters. As a result, the dominantfrequency component of the signals X_(A) and X_(B) remains at the inputfrequency ω_(in), which is then properly detected by the RSSI blocks.

A calibration clock used for the control logic runs at a frequency of250 kHz. The reference clock can be divided down inside the controller,or alternatively in the control logic. This clock frequency has beenselected to allow the RSSI outputs to settle after the capacitance valuein one of the polyphase filters has been incremented or decremented. Fora clock frequency of 250 kHz and a 4-bit control word generating 2⁴possible capacitance values, the calibration is completed within (250kHz)⁻¹(2⁴−1)=60 μs. During the calibration process the calibrationcircuitry draws 4 mA from a 3-V supply, and the RC calibration circuitrycan be powered down when the optimal RC value has been selected toreduce power consumption.

4.3 The Capacitor Array

In the transmitter, receiver and LO generator, metal-insulator-metal (M)capacitors can be used as the calibration component for the RC circuits.As those skilled in the art will appreciate, other capacitortechnologies may be used. The MIM capacitors are generally characterizedby a low bottom-plate parasitic capacitance to substrate of 1%.

A parallel capacitor array can be used in calibrating each RC circuit asshown in FIG. 41. The parallel array is much smaller in area than aseries array for the same capacitor value.

Complementary MOS switches or other switches known in the art, can beused in the capacitor array. The capacitor array can include any numberof capacitors. In the exemplary embodiment, the capacitor arraycapacitors 290, 292, 294, 296, 298 are connected in parallel. Switches300, 302, 304, 306 are used to switch the capacitors 292, 294, 296, 298,respectively, in and out of the capacitor array. In the describedembodiment, capacitor 290 is 2.4 pF, capacitor 292 is 2.4 pF, capacitor294 is 1.2 pF, capacitor 296 is 0.6 pF, capacitor 298 is 0.3 pF. Theswitch positions are nominally selected to produce an equivalentcapacitance equal to 4.8 pF. A code of “0111” means that capacitors 294,296, 298 are switched out of the capacitor array and capacitors 290, 292are in parallel.

The switches can be binary-weighted in size and the switch sizes can bechosen according to tradeoffs regarding parasitic capacitances andfrequency limitations based on the on-resistance of the CMOS switches.The capacitive error resulting from the parasitic capacitance in eachcapacitive array does not result in frequency error between the threepolyphase stages of the RC calibration circuit in the controller. Thisis because by using same capacitor array in each filter, and by scalingthe resistance accordingly in each case. Scaling resistances, relativeto those in the fundamental polyphase filter, by factors of ⅓ and ⅕ inthe 3^(rd) and 5^(th) harmonic filters respectively, are achieved with ahigh degree of accuracy with proper layout. Similarly, RC tuning in allother blocks utilizing the calibrated code is optimized when anidentical capacitive array is used, scaling only the resistance value intuning to the desired frequency. The capacitors in the capacitive arraysare laid out in 100 fF increments to improve the matching and parasiticfringing effects.

4.4 Bandgap Calibration Circuit for Accurate Bandgap Reference Current

In accordance with an exemplary embodiment of the present invention, abandgap reference current is generated by a bandgap calibration circuit.The bandgap reference current is used by the receiver, transmitter, andLO generator to set amplifier gains and voltage swings. The bandgapcalibration circuit generates an accurate voltage and resistance. Anaccurate bandgap reference current results from dividing the accuratevoltage by the an accurate resistance.

Bandgap calibration circuits can provide increased accuracy for improvedperformance. Embodiments of the present invention that are integratedonto a single IC can utilize bandgap calibration circuits to compensatefor process, temperature, and power supply variations. For example,variations in the absolute value of the resistance in a bandgapreference may result in deviations from optimal performance in sensitivecircuitry that rely on accurate biasing conditions. In the describedexemplary embodiment of the transceiver, a bandgap calibration circuitin the controller 16 provides an effective technique forself-calibration of resistance values in the transmitter, receiver andLO generator. The calibrated resistance values obtained as a result ofthe algorithm employed in the bandgap calibration circuit generate abias current that varies by only +2% over typical process, temperature,and supply variation.

Embodiments of the present invention which are integrated into a singleIC can use the described bandgap calibration circuit to provide accurateon-chip resistors by comparing the on-chip resistances to an off-chipreference resistor with a low tolerance of 1%. Using this method,trimming of on-chip resistance values with a total tolerance of 2% canbe achieved.

FIG. 42 shows an exemplary embodiment of the bandgap calibration circuitThe bandgap calibration circuit uses the reference clock provided fromthe LO generator and a reference resistor R_(REF) 236 to adjust atunable resistance value R_(POLY) 238 in a compare-and-increment loopuntil an optimum value is obtained. In embodiments of the presentinvention which are integrated into a single IC, the reference resistorR_(REF) 236 can be off-chip to provide improved calibration accuracy. A4-bit control word is output to accurately calibrate the resistors inthe transmitter, receiver and LO generator within ±2%. Transistors 227,226, 228, 230, 232, 234 form a cascode current with a reference currentI_(REF). The transistors 224, 230 each have their gates tied to theirrespective sources to set up the reference current I_(REF). By tying thegates of the transistors 224, 230, respectively to the gates of thetransistors 226,232, the reference current I_(REF) is mirrored to thereference resistor R_(REF 236). Similarly, by tying the gates of thetransistors 228, 234, respectively to the gates of the transistors, thereference current I_(REF) is also mirrored to the tunable resistorR_(POLY) 238. The voltage generated across the tunable resistor R_(POLY)238 is compared, using a latched comparator 240, to the voltagegenerated across the reference resistor R_(REF) 236. The value of thetunable resistor R_(POLY) 236 is incremented in successive steps,preferably, every 0.5 μs, through the utilization of control logic 242that is clocked, by way of example, at 2 MHz. This process continuesuntil the voltage V_(POLY) across the tunable resistor R_(POLY) 238matches the voltage V_(REF) across the off-chip reference resistorR_(REF) 236 causing the output of the comparator to change states anddisable the control logic 242. Once the control logic is disabled, the4-bit control word can be used to accurately calibrate the resistors inthe transmitter, receiver and LO generator.

The clock signals used by the calibration circuit are generated by firstdividing the reference clock input into the controller from the LOgenerator down in frequency, and then converting the result intodifferent phases for the comparison and increment phases of calibration.This bandgap calibration circuit provides accurate resistance values foruse in various on-chip circuit implementations because resistor scalingand matching on the same integrated circuit can be well controlled withproper layout techniques. The bandgap calibration circuit provides aresistor tuning range of approximately +30%, which is sufficient tocover the range of process variation typical in semiconductorfabrication. With a 4-bit control word generating 24 possible resistancevalues, the calibration is completed within (2 MHz)−1(24−1)=7.5 ms. Thecalibration circuit can be powered down when the optimal resistancevalue has been obtained.

The bandgap calibration circuit can be used for numerous applications.Byway of example, FIG. 43 shows a bandgap calibration circuit 244 usedin an application for calibrating a bandgap reference current that isindependent of temperature. The 4-bit control word from the bandgapcalibration circuit is coupled, by way of illustration, to the receiver.The 4-bit control word is used to calibrate resistances in aproportional-to-absolute-temperature (PTAT) bias circuit 246, and alsoin a V_(BE) (negative temperature coefficient) bias circuit 248. Theoutputs of these blocks are two bias voltages, V_(P) 250 and V_(N), 252that generate currents exhibiting a positive temperature coefficient,and a negative temperature coefficient, respectively. When thesecurrents are summed together using the cascode current mirror formed bytransistors 254, 256, 258, 260, the result is a current I_(OUT) displaysa (ideally) zero temperature coefficient

4.5 Resistor Array

In the transmitter, receiver and LO generator non-silicided polysiliconresistors can be used. As those skilled in the art will appreciate,other resistor technologies can also be used. Non-silicided polysiliconresistors have a high sheet resistance of 200-Ω/square along withdesirable matching properties. A switching resistor array as shown inFIG. 44 can be used to calibrate a resistor. The array includes serialconnected resistors 208, 210, 212, 214, 216, which, by way of example,have resistances of 2200Ω, 1100Ω, 550Ω, 275Ω, and 137Ω, respectively.The resistors 210, 212, 214, 216 include a bypass switch for switchingthe resistors in and out of the array. The switch positions arenominally selected to produce an equivalent of 3025 Ω. This resistancevalue has been chosen as a convenience to match the value used ingenerating an accurate bandgap reference current. A 4-bit calibrationcode 206 is used to control the total resistance in this array. As seenin FIG. 44, the resistances are binary-weighted in value and theaccurate scaling of each incremental resistance results by placing thelargest resistor (2200 Ω) 208 in series to generate each value. In thedescribed embodiment, the incremental resistances shown in FIG. 44 arechosen so that the total resistance in the array covers a range 30%above and below its nominal value, with a maximum resistance error of+2% determined by the incremental resistance switched by the LSB. Therange of resistance covered by the array is sufficient to cover typicalprocess variations in a semiconductor process. A series resistive arraymay be desirable opposed to a parallel resistive array because of thesmaller area occupied on the wafer.

CMOS switches are one of several different types of switch technologythat can be used. The sizing of the switches entails a tradeoff betweenthe on-resistance of each switch and the frequency limitations thatresult from the parasitic capacitances associated with each switch. Forcalibration resistors in the bandgap reference circuits, large switchesare used to minimize the effect of the on-resistance of each switch, asfrequency limitations are not a concern for this application.

5.0 Floating MOSFET Capacitors

Embodiments of the present invention that are integrated into a singleIC can be implemented with a variety of technologies including, by wayof example, CMOS technology. Heretofore, CMOS capacitors between twonodes with similar voltages (i.e., floating capacitors) have beenproblematic. In the described exemplary embodiment of the presentinvention, a MOS capacitor is used between two nodes having similarvoltages for signals with no DC information. The capacitor is made oftwo MOS capacitors in series with a large resistor in between to groundfor biasing.

FIG. 45 is a block diagram of the Floating MOS capacitor in accordancewith an embodiment of the present invention. As shown in FIG. 45, thecapacitor comprises two similar devices 858, 860 in series. Each MOStransistor has its source and drain connected together. The connecteddrain-source terminal of the MOS transistor 858 constitutes the input ofthe CMOS capacitor and the connected drain-source terminal of the MOStransistor 860 constitutes the output of the CMOS capacitor. The gatesof each MOS transistor are connected through a common resistor 862 to abias source (not shown).

6.0 Duplexing

In an alternative embodiment of the present invention, an integratedmatching circuit can be used to connect the LNA in the receiver to thePA in the transmitter. As the level of integration in radiocommunication circuits tend to grow, more functions are embodied on thesame chip and off-chip components are used less than ever. Presence ofexternal components not only augments the manufacturing costs, but alsoincreases the pin count on the main chip. The antenna switch is anexample of such components. This switch is used to connect the receiverto antenna in reception mode and the transmitter to antenna intransmission mode. In the described exemplary embodiment of the presentinvention, the antenna switch can be eliminated, and the input of thereceiver can be tied to the output of the transmitter. This approach hasvarious applications including, but not limited to, single chipintegration.

Since the antenna is usually single-ended, differential applicationsgenerally require a mechanism to convert the antenna signal fromsingle-ended to differential for connection to the differential lownoise amplifier (LNA) or the differential PA. The circuit implementationfor a single-ended to differential LNA is shown in FIGS. 46 and 47. LCcircuit, 646, 648 and the CL circuit 652, 650 matches the PA to theantenna when the PA is on and the LNA is off (as shown in FIG. 46), andmatches the LNA to the antenna when the LNA is on and the PA is off (asshown in FIG. 47). Since the LNA is off and it only introduces acapacitive loading to the PA. The matching circuit can be designed tocompensate for this additional capacitance.

In operation, during the transmit mode, a differential voltage acrossthe drains of the PA transistors 634, 632 is generated. The two drainsassert 180-degree out of phase voltages and they are combined throughthe LC and CL matching circuits to yield a single-ended voltage at theoutput. The LC circuit shifts the phase of the output signal from thetransistor 634 by 90 degrees. The CL circuit shifts the phase of thesignal output from the transistor 632 by 90 degrees in the oppositedirection. Consequently, both signals are in-phase when combined at theoutput of te matching circuits.

Although a preferred embodiment of the present invention has beendescribed, it should not be construed to limit the scope of the appendedclaims. For example, the present invention can be into a singleintegrated circuit, can be constructed from discrete components, or caninclude one or more integrated circuits supported by discretecomponents. Those skilled in the art will understand that variousmodifications may be made to the described embodiments. Moreover, tothose skilled in the various arts, the invention itself herein willsuggest solutions to other tasks and adaptations for other applications.It is therefore desired that the present embodiments be considered inall respects as illustrative and not restrictive, reference being madeto the appended claims rather than the foregoing description to indicatethe scope of the invention.

1. A method for providing a capacitance device in an integrated circuit,comprising: providing an input of the capacitance device, the input ofthe capacitance device comprising a first source node and a first drainnode of a first transistor; operatively coupling the first source nodewith the first drain node; operatively coupling a first gate node of thefirst transistor to a second gate node of a second transistor; providingan output of the capacitance device, the output of the capacitancedevice comprising a second source node and a second drain node of thesecond transistor; and operatively coupling the second source node tothe second drain node.
 2. The method according to claim 1, comprising:operatively coupling a resistor to the first gate node and the secondgate node.
 3. The method according to claim 1, wherein the firsttransistor is in series with the second transistor.
 4. The methodaccording to claim 1, comprising: operatively coupling a bias source tothe first gate node and the second gate node.
 5. The method according toclaim 1, comprising: inputting a signal at the input of the capacitancedevice.
 6. The method according to claim 5, wherein the signal carriessubstantially no DC information.
 7. The method according to claim 1,comprising: outputting a signal at the output of the capacitance device.8. The method according to claim 7, wherein the signal carriessubstantially no DC information.
 9. The method according to claim 1,wherein the first transistor and the second transistor use CMOStechnology.
 10. The method according to claim 1, wherein the capacitancedevice comprises a floating capacitance device.
 11. A method forproviding a capacitance device, comprising: (a) operatively coupling afirst transistor to a second transistor in series; (b) inputting aninput signal to an input of the first transistor, the input of the firsttransistor comprising a first node and a second node of the firsttransistor; (c) operatively coupling the first node and the second nodeof the first transistor; (d) operatively coupling a third node and afourth node of the second transistor; and (e) outputting an outputsignal from an output of the second transistor, the output of the secondtransistor comprising the third node and the fourth node.
 12. The methodaccording to claim 11, wherein (a) comprises operatively coupling afifth node of the first transistor to a sixth node of the secondtransistor.
 13. The method according to claim 12, wherein the fifth nodecomprises a gate node of the first transistor, and wherein the sixthnode comprises a gate node of the second transistor.
 14. The methodaccording to claim 11, comprising: integrating the capacitance deviceinto an integrated circuit.
 15. The method according to claim 11,wherein the first node of the first transistor comprises a source nodeof the first transistor, wherein the second node of the first transistorcomprises a drain node of the first transistor, wherein the third nodeof the second transistor comprises a source node of the secondtransistor, and wherein the fourth node of the second transistorcomprises a drain node of the second transistor.
 16. The methodaccording to claim 11, comprising; (f) biasing the first transistor andthe second transistor via the fifth node of the first transistor and thesixth node of the second transistor.
 17. The method according to claim16, wherein (f) comprises biasing the first transistor and the secondtransistor via a shared resistor.
 18. The method according to claim 11,wherein the capacitance device provides a floating capacitance.
 19. Themethod according to claim 11, wherein the input signal carriessubstantially no DC information, and wherein the output signal carriessubstantially no DC information.
 20. The method according to claim 11,wherein the first transistor or the second transistor uses MOStechnology.